Method and circuit arrangement for regulating a LED current flowing through a LED circuit arrangement, and associated circuit composition and lighting system

ABSTRACT

Consistent with an example embodiment, there is a method for regulating a LED current (ILED) flowing through a LED circuit arrangement at a mean LED current level. The method includes establishing an oscillating converter current (IL), establishing a first and a second current control indicator representative of a flow of a converter current (IL); regulating a peak and valley current level of the converter current in dependence on the first current control indicator; controlling a converter current period (T) of an oscillation of the converter current in dependence on the second current control indicator to be within a period control range (Tref) and feeding at least part of the converter current to the LED circuit arrangement.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of copending U.S. patent applicationSer. No. 12/864,803, filed on Jul. 27, 2010, which is a 371 ofinternational patent application PCT/IB2009/050342, filed on Jan. 28,2009, which claims the benefit of EP patent application no. 08101113.2,filed on Jan. 30, 2008.

TECHNICAL FIELD

The invention relates to a method regulating a LED current flowingthrough a LED circuit arrangement at a mean LED current level. Theinvention further relates to a circuit arrangement for regulating a LEDcurrent flowing through a LED circuit arrangement at a mean LED currentlevel. The invention further relates to a LED driver IC. The inventionfurther relates to a circuit composition and to a LED lighting system.

BACKGROUND

The light output of a light emitting diode (LED) is generally controlledby regulating a current level of a LED current through the LED. The LEDcurrent may be further modulated with, e.g. a pulse width modulation(PWM) scheme. In such a PWM-scheme, the LED receives the LED current ina periodic sequences of pulses of a certain width, while the width ofthe pulses is modulated from a first pulse width to a second pulse widthwhen the effective light output is to be changed from a first lightoutput level to a second light output level.

A LED drive method and a LED drive circuit thus generally comprise acurrent source, providing a constant current or an oscillating currentwith an average current level, and a switch associated with the LED inorder to control a path of the current and in order to achieve the pulsewidth modulation of the LED current.

The switch may be in series with the LED, thus controlling the path ofthe current by interrupting the path of the current in order to achievethe pulse width modulation.

The switch may alternatively be in parallel with the LED, which will bereferred to as a bypass switch. The bypass switch controls the path ofthe current by either guiding the path of the current through the LED orguiding the path of the current through a bypass path parallel to theLED in order to achieve the pulse width modulation. One of theadvantages of such a bypass switch approach is that the currentcontinues to flow, either through the LED or though the bypass path,which allows the use of very efficient current sources, such as aswitch-mode current source. This is especially advantageous when aplurality of LEDs are to be operated at a common current level but witha possibly different pulse width between different LEDs from theplurality of LEDs. The LEDs may then be arranged in a plurality of LEDsegments connected in series, each LED segment comprising a single LEDor two or more LEDs, the two or more LEDs preferably arranged in series,and each of the LED segments being associated with a bypass switch inparallel to the corresponding LED segment. By operating the bypassswitches independently, the effective light output of each of the LEDsegments may be varied independently.

An example of a current source is described in WO2004100614A1.WO2004100614A1 describes a LED current control method and circuit foraccurately and quickly regulating the mean amperage of LED currentduring all operating conditions including a change in the input line ofa power source or in a change in a load of the LED network.

The method comprises controlling the LED current to oscillate, e.g. in atriangular or saw-tooth manner, between a peak amperage and a valleyamperage, with the mean amperage being the average of the peak amperageand the valley amperage, by an alternate controlling of an increase anda decrease of the LED current in response to each crossover by aconverter current sensing voltage of a lower trip voltage and an uppertrip voltage in a negative and a positive direction respectively. Acircuit using such a method may be referred to as an example of aswitch-mode converter with hysteretic control on the LED current. Thepeak-to-valley range of the peak amperage to the valley amperage may bereferred to as the hysteretic current window. The peak-to-valley rangeof the upper trip voltage to the lower trip voltage may be referred toas the hysteretic voltage window, or, in short, the hysteretic window.

The method and circuit thus achieve regulating the mean current levelindependent of the operating conditions. In particular, when the methodand circuit are used to operate a LED circuit arrangement comprising aplurality of LED segments with corresponding bypass switches in anarrangement as described above. Operating the bypass switches to varythe light output of the individual LED segments results in a variationof the load of the LED circuit arrangement. The switch-mode converterwith hysteretic control is well suited to accurately and quickly delivera current with a substantially constant mean current level to such a LEDcircuit arrangement with varying load due to the operation of the bypassswitches.

However, using the method of WO2004100614A1 results in a varyingfrequency of the oscillation of the LED current when the load of the LEDcircuit arrangement is varying, e.g. due to the operation of the bypassswitches as described above. When the load of the LED circuitarrangement is varying significantly, the frequency variation may belarge.

This large variation of frequency has several negative side-effects. Forexample, the components in an input or output filter of the hystereticswitch-mode converter need to be dimensioned such as to reduce theside-effects to a sufficiently low level for all possible frequencies.The requirements stemming from these side-effects are for examplepreventing audible noise, preventing visible and potentially annoyingfluctuations in the light output of the LEDs, complying to conducted andradiated electromagnetic interference (EMI) regulations, guaranteeinglifetime of electrolytic capacitors, and optimizing core versusconduction losses in inductors. For example, in order to guarantee thatthe frequency does not, during all operating conditions, move into therange of audible frequencies, which may be annoying, a small inductor isneeded in the output filter, which negatively impacts the accuracy ofthe LED current level. As another example, in order to achieve a smallripple of the LED current when the load of the LED circuit arrangementis varying significantly while being tolerant to changes in the inputline of the power source, a large capacitor may be needed to filter theinput voltage.

SUMMARY

The present invention aims to reduce the side-effects of the knownswitch-mode converters with hysteretic control and aims to provide acircuit arrangement and a method which can handle a significant loadvariation of the LED network with reduced negative side-effects.

For this purpose, the method according to the invention comprises:

-   -   establishing a converter current;    -   establishing a first current control indicator representative of        a current level of the converter current;    -   establishing an oscillation of the converter current between a        valley current level and a peak current level in dependence on        at least the first current control indicator, wherein the mean        LED current level corresponds to a weighted average of the peak        current level and the valley current level of the converter        current;    -   establishing a second current control indicator representative        of a flow of the converter current;    -   controlling a converter current period of the oscillation of the        converter current to be within a period control range, the        controlling being performed in dependence on at least the second        current control indicator, and    -   feeding the LED circuit arrangement with at least part of the        converter current.

The first and second current control indicator may be a signal, e.g.running over an electrical connection between two components of acircuit arrangement. or a value, e.g. a value of a parameter stored in aregister or a memory.

The mean current level of the LED current may, e.g. be regulated in amanner similar to that of the prior-art switch-mode converter withhysteretic control, as may, e.g. be applied for a Buck-converter feedinga LED circuit arrangement of a plurality of LEDs in a series arrangementwith bypass switches in parallel to each of the LEDs. The hystereticcontrol is applied on the converter current. For such a converter, theconverter current behaves as a continuous, typically sawtooth-shapedcurrent and the full converter current established in the converter isfed as the LED current to the LED circuit arrangement with the mean LEDcurrent level corresponding to the arithmetic average of the peakcurrent level and the valley current level. When the LED circuitarrangement comprises a capacitive filter in parallel to the seriesarrangement of the plurality of LEDs, the LED current behaves as atime-filtered version of the converter current. The peak and valleyvalues of the LED current may then differ from the respective values ofthe converter current, while the mean LED current level may still levelcorrespond to the arithmetic average of the peak current level and thevalley current level. For different types of converters, such as forexample a Buck-Boost converter, the LED current may be discontinuouseven when the converter current is continuous: the hysteretic controlmay then be performed on the converter current, and part of theconverter current will be fed to the LED circuit arrangement as the LEDcurrent. The mean LED current level may then correspond to a weightedaverage of the peak current level and the valley current level withdifferent weights for the peak current level and the valley currentlevel, to take the effects if the partial feeding into account.

It should be remarked that the method according to the inventionmonitors and controls the converter current, whereas the method ofWO2004100614A1 uses the LED current. These currents are the same for aBuck converter feeding a LED circuit arrangement of a series arrangementof a plurality of LEDs, but may be different for other types ofconverters, e.g. for a Buck-Boost converter which may be arranged,depending on its implementation, to feed the LED circuit arrangementonly during the part of the converter current period during which theconverter current is increasing or only during the part of the convertercurrent period during which the converter current is decreasing. Forthose types of converters, hysteretic control is preferably performed onthe converter current.

The period control range may be a narrow window around a pre-determinedset-point frequency, in order to achieve a substantially constantduration of the converter current period.

The period control range may alternatively be, e.g. a window between apre-determined lower threshold duration and a pre-determined upperthreshold duration, wherein the lower and upper thresholds duration may,e.g. be associated with preferred values of electrical components in thecircuit arrangement or electrical components connected to the circuitarrangement, such as capacitors or inductors. Alternatively, the lowerand upper thresholds duration may, e.g. be associated with a preferredfrequency range, such as a frequency range excluding audiblefrequencies, a frequency range associated with specific electromagneticinterference risks, or a frequency range excluding annoying periodicfluctuations in the light output level, known as flicker.

The period control range may have a fixed value, which may be apre-determined constant value. The period control range mayalternatively be adjusted during operation, e.g. to spread the spectralenergy associated with the operation over a spectral band in order tomeet practical or legal electromagnetic compatibility requirements.

Controlling the converter current period may be performed by a varietyof different methods, employing, e.g. a direct determination of theconverter current period duration such as a measurement on an electricalsignal associated with the circuit arrangement, or, e.g. an indirectdetermination of the converter current period duration. Embodiments forcontrolling the converter current period are described below.

In an embodiment of the method,

-   -   establishing the first current control indicator comprises:        -   monitoring a current level of the converter current and            using the monitored current level as the first current            control indicator;    -   establishing the oscillation of the converter current comprises:        -   establishing an upper trip current level and a lower trip            current level as control crossover thresholds, the upper            trip current level being associated with the peak current            level of the converter current and the lower trip current            level being associated with the valley current level of the            converter current;        -   controlling an increase of the converter current from a            valley current level to a peak current level in response to            each crossover of the lower trip current level by the            current level of the converter current in a negative            direction, the controlling of the increase of the converter            current being associated with an increase time duration, and        -   controlling a decrease of the converter current from the            peak current level to the valley current level in response            to each crossover of the upper trip current level by the            current level of the converter current in a positive            direction, the controlling of the decrease of the converter            current being associated with a decrease time duration, and    -   in controlling the converter current period of the oscillation        of the converter current, the converter current period        corresponds to a sum of the increase time duration and the        decrease time duration.

The mean current level of the LED current is thus regulated in a mannersimilar to that of the prior-art switch-mode converter with hystereticcontrol. The method controls the mean LED current level by controllingthe converter current to oscillate between a peak current level and avalley current level in response to each crossover by the convertercurrent of the lower trip current level and the upper trip current levelin the negative and the positive direction respectively. The lower tripcurrent level and the upper trip current level may be established independence on the required mean current level of the LED current.

The oscillation comprises alternated control periods with the increasetime duration in which the converter current is increased and with thedecrease time duration in which the converter current is decreased.

The invention provides a new inventive aspect to the switch-modeconverter with hysteretic control in providing additionally a control ofthe converter current period, or, equivalently the frequency, of theoscillation of the converter current. The converter current period iscontrolled to be within a range, referred to as the period controlrange.

In a further embodiment of the method,

-   -   establishing the first current control indicator comprises:        -   establishing a converter current sensing voltage            representative of a flow of the converter current through            the LED circuit arrangement;    -   regulating the mean current level of the converter current        comprises        -   establishing an upper trip voltage and a lower trip voltage            as control crossover thresholds, the upper trip voltage            (being associated with the peak current level of the            converter current, the lower trip voltage being associated            with the valley current level of the converter current, and            the mean current level being an average of the peak current            level and the valley current level of the converter current;        -   controlling an increase of the converter current from the            valley current level to the peak current level in response            to each crossover of the lower trip voltage by the converter            current sensing voltage in a negative direction, the            controlling of the increase of the converter current being            associated with an increase time duration, and        -   controlling a decrease of the converter current from the            peak current level to the valley current level in response            to each crossover of the upper trip voltage by the converter            current sensing voltage in a positive direction, the            controlling of the decrease of the converter current being            associated with a decrease time duration, and    -   in controlling the converter current period of the oscillation        of the converter current, the converter current period        corresponds to a sum of the increase time duration and the        decrease time duration.

The current level of the converter current is thus represented by theconverter current sensing voltage, allowing an easier electrical signalmanipulation and signal processing than a current signal. The convertercurrent sensing voltage may, e.g. be the voltage over a resistor in thepath of the converter current.

The method may further comprise

-   -   determining an adjusted upper trip voltage value and an adjusted        lower trip voltage value in dependence with the converter        current period,    -   establishing the upper trip voltage and the lower trip voltage        from the adjusted upper trip voltage value and the adjusted        lower trip voltage value, and    -   wherein controlling the converter current period to be within        the period control range is associated with using the adjusted        upper trip voltage value) and the adjusted lower trip voltage in        controlling the increase of the converter current and        controlling the decrease of the converter current.

The upper trip voltage and the lower trip voltage can have differentvalues for achieving the same mean current level, as the mean currentlevel may be the same for different pairs of peak current level andvalley current level.

However, each choice for a specific upper trip voltage and a specificlower trip voltage, associated with a specific peak current level and aspecific valley current level, results in a specific, and generallydifferent, converter current period when the current level is regulatedwithout controlling the converter current period. By adjusting the uppertrip voltage and the lower trip voltage while maintaining the meancurrent level, the associated decrease time duration and increase timeduration can be adjusted. Hence the converter current period can becontrolled by adjusting the upper trip voltage and the lower tripvoltage. One may note that adjusting the upper trip voltage and thelower trip voltage may result in an adjustment of the peak current leveland valley current level, at the same mean current level.

Controlling may be based on the observation that the converter currentperiod will become smaller when the difference between the upper tripvoltage value and the lower trip voltage becomes smaller, while theconverter current period will become larger when the difference becomeslarger, as the speed of increase and the speed of decrease of theconverter current are unchanged when the load of the LED circuitarrangement is unchanged. Controlling can thus be performed in a defineddirection with, e.g. predetermined step sizes of the adjustments to theupper trip voltage and the lower trip voltage, the adjustments beingdone to the upper trip voltage and the lower trip voltage in oppositedirections. This allows, e.g. the use of a feed-back control loop ofwhich a variety of practical implementations will be familiar to aperson skilled in the art.

In an embodiment, determining the adjusted upper trip voltage and theadjusted lower trip voltage comprises retrieving at least onevoltage-related value from a memory.

The memory may, e.g. comprise pre-determined values for the upper tripvoltage and the lower trip voltage, or one or two adjustment values. Theat least one voltage value may, e.g. be organized in the memory as afunction of a converter current period, a load of the LED circuitarrangement, or any other parameters associated with the convertercurrent period associated with specific values of the upper trip voltageand the lower trip voltage.

In an embodiment, the adjusted upper trip voltage and the adjusted lowertrip voltage are determined from the upper trip voltage, the lower tripvoltage and an adjustment voltage.

The adjustment voltage may relate to the difference between the adjustedupper trip voltage and the adjusted lower trip voltage.

Use of a single adjustment voltage has the advantage that in determiningthe adjusted upper trip voltage and the adjusted lower trip voltage, theaverage of these two voltages may be kept unchanged, e.g. by adding halfthe adjustment voltage to the upper trip voltage to achieve the adjustedupper trip voltage and by subtracting half the adjustment voltage fromthe lower trip voltage to achieve the adjusted lower trip voltage, as:VHA=VH+VADJ/2,VLA=VL−VADJ/2,wherein VH denotes the upper trip voltage, VL denotes the lower tripvoltage, VHA denotes the adjusted upper trip voltage, VLA denotes theadjusted lower trip voltage and VADJ denotes the adjustment voltage.

In an embodiment, the adjusted upper trip voltage and the adjusted lowertrip voltage are determined from a reference voltage and an hysteresisvoltage.

Use of a reference voltage and an hysteresis voltage has the advantagethat in determining the adjusted upper trip voltage and the adjustedlower trip voltage, the average of these two voltages, relating to themean current level, can be determined from the reference voltage, suchthat an additional change in the mean current level can be easilyincorporated in the controlling. The adjusted upper trip voltage and theadjusted lower trip voltage may then be determined by adding half thehysteresis voltage to the reference voltage to achieve the adjustedupper trip voltage and by subtracting half the hysteresis voltage fromthe reference voltage to achieve the adjusted lower trip voltage, as:VHA=VREF+VHYS/2,VLA=VREF−VHYS/2,wherein VHA denotes the adjusted upper trip voltage, VLA denotes theadjusted lower trip voltage, VREF denotes the reference voltage and VHYSdenotes the hysteresis voltage.

In an embodiment, establishing the second current control indicatorcomprises measuring a converter current period duration of the convertercurrent period, and the measured converter current period duration isused as the second current control indicator in controlling theconverter current period.

Controlling the converter current period using the measured convertercurrent period duration, as a feed-back control, has the advantage thata very accurate control may be achieved, as not only all known and/oranticipated effects can be incorporated in the controlling, but alsoother effects, e.g. accidental, occasional or ageing-related effects,are incorporated.

Measuring the converter current period duration may comprise acquiring aplurality of converter current period duration values and filtering theacquired values to obtain a measured converter current period durationthat is substantially free from noise.

Measuring the converter current period duration may be performed withthe converter current or the converter current sensing voltage as ameasurement signal.

Measuring the converter current period duration may alternatively beperformed with a control signal associated with controlling an increaseand the decrease of the converter current as the measurement signal.

Measuring the converter current period duration may be performed using,e.g. analyzing a period, analyzing a frequency or analyzing a spectrum.

Measuring may comprise sampling the measurement signal with ahigh-frequency sampling clock, e.g. a system clock of a processor unitor a clock derived there from.

In an alternative embodiment, establishing the second current controlindicator comprises determining a load associated with the LED circuitarrangement, and the load is used the second current control indicatorin controlling the converter current period. In controlling theconverter current period using the load, the load may be associated withan estimate of the converter current period.

The load associated with the LED circuit arrangement may be determinedfrom, e.g. the operating conditions of the LED circuit arrangement. Asan example, in the LED circuit arrangement comprising a plurality of LEDsegments with corresponding bypass switches as discussed above, thestate of the plurality of bypass switches determines the load of the LEDcircuit arrangement. The state of the plurality of bypass switches maythus be used to estimate the load. Controlling the converter currentperiod may then, e.g. comprise determining the adjusted upper tripvoltage values and the adjusted lower trip voltage values from a smallpre-determined table comprising the adjusted upper trip voltage valuesand the adjusted lower trip voltage values associated with thepredetermine window of the converter current period as a function of thestate of the plurality of bypass switches.

Controlling the converter current period in a feed-forward manner usingthe estimated converter current period duration has the advantage that asimple control may be achieved with a still sufficient accuracy.

Alternatively, the load associated with the LED circuit arrangement maybe measured from the LED circuit arrangement. As an example, the voltagedrop over the LED circuit arrangement may be associated with the load ofthe LED circuit arrangement. Measuring this voltage drop is thussuitable for determining the load. As an alternative example, signallevels of the signals controlling the bypass switches may be measuredand used to determined the load.

In a further embodiment, establishing the second current controlindicator comprises determining an estimate of a converter currentperiod duration of the converter current period from the load, and theestimate of the converter current period duration is used as the secondcurrent control indicator in controlling the converter current period.

In an embodiment,

-   -   the period control range is defined with a lower period        threshold and an upper period threshold, and    -   the lower duration threshold and the upper duration threshold        are determined from a centre duration and a duration width,        wherein the duration width is smaller than 10% of the centre        duration.

The converter current period is thus controlled within a well-definedwindow.

The centre duration thus corresponds to centre frequency, and theduration width corresponds to a frequency band around this centrefrequency.

In a further embodiment, the centre duration has a constant value.

The converter current period is thus controlled to remain at asubstantially constant, fixed duration.

In an alternative further embodiment, the centre duration is varied overa spectral band.

The converter current period is thus controlled in a spread spectrummanner, to prevent a concentration of the spectral energy bydistributing the spectral energy over a spectral band, e.g. in order tomeet practical or legal electromagnetic compatibility requirements witha reduced amount of electromagnetic interference (EMI) filtering therebyachieving a cost advantage.

In embodiments of the method, the method further comprises:

-   -   controlling a path of the LED current flowing through the LED        circuit arrangement,    -   wherein the LED circuit arrangement comprises a first LED        segment and at least a second LED segment, the first LED segment        being associated with a first switching element operable for        controlling a path of the LED current through the first LED        segment, the second LED segment being associated with a second        switching element operable for controlling a path of the LED        current through the second LED segment.

Controlling the path of the LED current flowing through the LED circuitarrangement by operating the first and the second switching element forcontrolling the path of the current through the first and the second LEDsegments is associated with varying the load of the LED circuitarrangement. The effects of this load variation may be effectivelydiminished by controlling the converter current period using the methodaccording to the invention.

In an embodiment of the method:

-   -   the first switching element is electrically parallel to the        first LED segment,    -   the second switching element is electrically parallel to the        second LED segment, and    -   the first and second switching elements are operated to select        the path of the LED current to pass through the LED segment        associated with the switching element or to bypass the LED        segment associated with the switching element.

When the first switching element is open, the current will flow throughthe first LED segment. When the first switching element is closed, thecurrent will flow through the first switching element and bypass thefirst LED segment.

When the second switching element is open, the current will flow throughthe second LED segment. When the second switching element is closed, thecurrent will flow through the second switching element and bypass thesecond LED segment.

By operating the first and second switching elements, the path of theLED current is thus selected to pass selectively through the LEDsegments.

In an embodiment, the load is derived from a status of the first andsecond switching elements.

Determining the adjusted upper trip voltage value and the adjusted lowertrip value may thus be performed from the status of the first and secondswitching elements without a direct measuring of the converter currentperiod.

An embodiment of the method comprises

-   -   storing a hysteresis voltage for the status of the first and        second switching elements in a memory unit before a change of        status of at least one of the first and second switching        elements, and    -   retrieving the hysteresis voltage for the status of the first        and second switching elements from the memory unit after a        change of status of at least one of the first and second        switching elements.

This allows to memorize e.g. the adjusted upper trip voltage, theadjusted lower trip voltage, the upper trip voltage, the lower tripvoltage, the adjustment voltage, the hysteresis voltage, after the LEDcircuit arrangement has been operated with a first load condition duringwhich the voltage has been accurately determined in, e.g. a controlemploying a feedback manner based on measuring the converter currentperiod as described above. When the LED circuit arrangement is then at alater moment in time again set to operate with the first load condition,the memorized value can be retrieved, thus improving the speed ofconvergence of the feedback control.

The circuit arrangement according to the invention provides a circuitarrangement for regulating a LED current flowing through a LED circuitarrangement at a mean LED current level, the circuit being arranged for:

-   -   establishing a converter current;    -   establishing a first current control indicator representative of        current level of the converter current through the circuit        arrangement;    -   establishing an oscillation of the converter current between a        valley current level and a peak current level in dependence on        at least the first current control indicator, wherein the mean        LED current level corresponds to a weighted average of the peak        current level and the valley current level of the converter        current;    -   establishing a second current control indicator representative        of a flow of the converter current through the circuit        arrangement;    -   controlling a converter current period of the oscillation of the        converter current in dependence on the second current control        indicator to be within a period control range, and    -   feeding the LED circuit arrangement with at least part of the        converter current.

The circuit arrangement can be used to implement one of the methoddescribed above in detail. The circuit arrangement may, during use, bein electrical connection to a LED circuit arrangement and may cooperatewith the LED circuit arrangement. The LED circuit arrangement mayalternatively be included in the circuit arrangement. Embodiments of thecircuit arrangement are described below.

In an embodiment of the circuit arrangement according to the invention:

-   -   for establishing the first current control indicator, the        circuit arrangement comprises:        -   a converter current sensor operable to establish a converter            current sensing voltage representative of the converter            current flowing through the LED circuit arrangement;    -   for regulating the mean current level of the converter current,        the circuit arrangement comprises:        -   a hysteretic comparator operable to establish an upper trip            voltage and a lower trip voltage as control crossover            thresholds, the upper trip voltage being associated with a            peak current level of the converter current, the lower trip            voltage being associated a valley current level of the            converter current, the hysteretic comparator in electrical            communication with the converter current sensor to receive            the converter current sensing voltage,        -   wherein the hysteretic comparator is operable to output a            switching control voltage at a first logic level in response            to each crossover of the lower trip voltage by the converter            current sensing voltage in a negative direction, and        -   wherein the hysteretic comparator is operable to output a            switching control voltage at a second logic level in            response to each crossover of the upper trip voltage by the            converter current sensing voltage in a positive direction,            and        -   a switch-mode converter operable to control a flow of the            converter current through the circuit arrangement, the            switch-mode converter in electrical communication with the            hysteretic comparator to receive the switching control            voltage,        -   wherein the switch-mode converter controls an increase of            the converter current from the valley current level to the            peak current level in response to the switching control            voltage equaling the first logic level, the controlling of            the increase of the converter current being associated with            an increase time duration, and        -   wherein the switch-mode converter controls a decrease of the            converter current from the peak current level to the valley            current level in response to the switching control voltage            equalling the second logic level, the controlling of the            decrease of the converter current being associated with a            decrease time duration, and    -   for controlling the converter current period of the oscillation        of the converter current, the circuit arrangement comprises:        -   a converter current period controller operable to control            the converter current period, the converter current period            controller being in electrical communication with at least            the hysteretic comparator;        -   wherein the converter current period corresponds to a sum of            the increase time duration and the decrease time duration.

The converter current period controller thus cooperates with theconverter current sensor, the hysteretic comparator and the switch-modeconverter to regulate the mean LED current level as well as theconverter current period.

The converter current sensor may comprise a resistor in the current pathof the converter current and a voltage measurement unit arranged tomeasure the voltage over the resistor and to provide the measuredvoltage as the converter current sensing voltage.

The converter current sensor may alternatively cooperate with a resistorin the current path of the converter current and comprise a voltagemeasurement unit arranged to measure the voltage over the resistor andto provide the measured voltage as the converter current sensingvoltage. The resistor may be a resistor external to the circuitarrangement but connected to the circuit arrangement. E.g. when thecircuit arrangement is an integrated circuit, the resistor may beconnected to the IC, and the IC may comprise the voltage meter tomeasure the voltage over the resistor.

In an embodiment, the converter current period controller comprises:

-   -   a trip control voltage generator operable to establish a first        trip control voltage and a second trip control voltage; and    -   the hysteretic comparator is operable to establish the upper        trip voltage and the lower trip voltage from the first trip        control voltage and the second trip control voltage, the        hysteretic comparator being in electrical communication with the        trip control voltage generator to receive the first trip control        voltage and the second trip control voltage.

As described above, by adjusting the upper trip voltage and the lowertrip voltage while maintaining the mean current level, the associateddecrease time duration and increase time duration can be adjusted. Hencethe converter current period can be controlled by adjusting the uppertrip voltage and the lower trip voltage.

In an embodiment, the converter current period controller comprises:

-   -   a converter current period detector operable to establish the        second current control indicator, the second current control        indicator being associated with the converter current period,        the converter current period detector being in electrical        communication with at least one of the group consisting of the        converter current sensor, the hysteretic comparator and a LED        circuit load detector, to receive at least one of the group        consisting of the converter current sensing voltage, the        switching control voltage and a LED circuit load,    -   wherein the trip control voltage generator is operable to        establish the first trip control voltage and the second trip        control voltage in response to the second current control        indicator associated with the converter current period in order        to control the converter current period to be within a period        control range, the trip control voltage generator being in        electrical communication with the converter current period        detector to receive the second current control indicator.

The converter current period detector may be in electrical communicationwith the converter current sensor or the hysteretic comparator toreceive the converter current sensing voltage. The converter currentperiod detector may be operable to determine the converter currentperiod duration from the converter current sensing voltage, e.g. bymeasuring the duration of the converter current period. The convertercurrent period duration may then be used as the second current controlindicator.

The converter current period detector may be in electrical communicationwith the hysteretic comparator to receive the switching control voltage.The converter current period detector may be operable to determine theconverter current period duration from the switching control voltage.The converter current period duration may then be used as the secondcurrent control indicator.

The converter current period detector may be in electrical communicationwith the LED circuit load detector to receive the LED circuit load. TheLED circuit load may then be directly used as the indicator.Alternatively, the LED circuit load may be used to estimate theconverter current period duration, and the estimated converter currentperiod duration may then be used as the second current controlindicator.

The converter current period detector may fully or in part beimplemented in a microprocessor in communication with the circuitarrangement.

In a further embodiment, the trip control voltage generator is operableto establish the first trip control voltage and the second trip controlvoltage from a reference voltage and a hysteresis voltage, and

-   -   the trip control voltage generator is operable to control the        hysteresis voltage in response to the second current control        indicator in order to control the converter current period.

The hysteretic comparator is operable to establish the upper tripvoltage and the lower trip voltage from the first trip control voltageand the second trip control voltage.

The reference voltage may relate to the mean current level and may bekept constant when the converter current period is controlled whilekeeping the mean current level unchanged.

The hysteresis voltage may relate to the oscillation of the current, andmay be adjusted in order to control the converter current period.

Controlling the hysteresis voltage may also be referred to ascontrolling the hysteresis window or as controlling the hysteresisvoltage window.

The first trip control voltage and the second trip control voltage maythen be determined by adding half the hysteresis voltage to thereference voltage to achieve the first trip control voltage and bysubtracting half the hysteresis voltage from the reference voltage toachieve the second trip control voltage, as:VC1=VREF+VHYS/2,VC2=VREF−VHYS/2,wherein VC1 denotes the first trip control voltage, VC2 denotes thesecond trip control voltage, VREF denotes the reference voltage and VHYSdenotes the hysteresis voltage.

In embodiments of the circuit arrangement, the circuit arrangementcomprises a resistive digital-to-analogue converter,

-   -   the resistive digital-to-analogue converter comprising:        -   a converter reference voltage supply arranged to provide a            converter reference voltage,        -   a series circuit of resistors in electrical communication            with the converter reference voltage supply, and        -   a first and a second switch array comprising a plurality of            switches,        -   wherein each of the switches of the first and second switch            array is in electrical communication with the series circuit            of resistors tapping off at a corresponding position along            the series circuit of resistors, and        -   wherein each switch array is arranged to be controlled with            a digital control word comprising a plurality of bits, the            bits being associated with controlling the switches to tap            off the series circuit of resistors at the corresponding            position, and        -   wherein the first trip control voltage and the second trip            control voltage are tapped from the same series circuit of            resistors with the first switch array and the second switch            array respectively.

The resistive digital-to-analogue converter (R-DAC) thus provides stableand well-defined voltages.

By providing two switch arrays tapping of the same series circuit ofresistors, a good correspondence is achieved between the two voltagestapped from the R-DAC.

The resistors in the series circuit of resistors may all have equalresistances.

Denoting the number of bits with B, the series circuit of resistors may,e.g. comprise a series arrangement of 2^(B-1) resistors, all resistorshaving an equal resistance, and each of the switch arrays may comprise2^(B) switches. Such an inherent structure of the R-DAC guarantees thatthe average between two voltages tapped of the R-DAC remains the samewhen stepping up and down with the same value, i.e. with the same numberof resistors.

As an example, when using an R-DAC with 1023 resistors, i.e. B=10, thedigital control word is a 10-bit word with codes values between 0 and1023. A situation may occur where for a first load of the LED circuitarrangement, the first trip control voltage corresponds to a code valueof 814, the second trip control voltage corresponds to a code value of776 and the converter current period has a first duration within theperiod control range. At a second load of the LED circuit arrangement,only slightly different from the first load, the method according to theinvention controls the converter current period to remain within theperiod control range and to substantially maintain the first duration,e.g. by stepping the first control voltage up with one unit in the codevalues and the second control voltage down with one unit in the codevalues. When, in this example, the second load is only slightlydifferent from the first load, this may be performed by adjusting thefirst trip control voltage to a new code value of 815 and the secondtrip control voltage corresponds to a new code value of 775.

A further embodiment comprises a R-DAC memory, wherein the digitalcontrol word of the R-DAC switch settings are stored in and retrievedfrom the R-DAC memory.

A setting may be stored when the load changes from a first load to asecond load, in order to memorize the setting for the first load. Whenat a later moment in time the load returns back to the first load, thememorized setting can be retrieved. As a result, the settling time forthe controlling will be reduced.

The R-DAC memory may be comprised with the R-DAC; when the R-DAC isimplemented in an integrated circuit, the R-DAC memory may beimplemented in the same integrated circuit.

The R-DAC memory may also be comprised in an external memory, e.g. in amemory associated with a microprocessor in communication with thecircuit arrangement.

In an embodiment of the circuit arrangement

-   -   the circuit arrangement comprises a memory unit comprising a        table comprising voltage settings for a plurality of indicator        values, and    -   the trip control voltage generator is operable to retrieve        voltage settings in response to the second current control        indicator in order to establish the first trip control voltage        and the second trip control voltage.

The memory unit may be pre-loaded with the table comprising voltagesettings for the plurality of indicator values.

The memory may be updated during operation of the circuit arrangement,e.g. when controlling the converter current period resulted in a voltagesetting deviating from the voltage setting retrieved from the table forone of the indicator values.

For example, the memory may comprise a table of adjustment voltagevalues for a plurality of load values, or a table of upper trip voltagevalues and lower trip voltage values for a plurality of load values.

An embodiment comprises a spread-spectrum generator arranged for varyinga centre duration of the period control range over a pre-determinedspectral band.

This allows the converter current period to be controlled in a manner toprevent a concentration of the spectral energy by distributing thespectral energy over a spectral band, e.g. in order to meet practical orlegal electromagnetic compatibility requirements.

In an embodiment of the circuit arrangement,

-   -   the hysteretic comparator comprises a comparator having an        inverting input and a non-inverting input,    -   the converter current sensing voltage is applied to the        inverting input of the comparator, and    -   the hysteretic comparator comprises a multiplexer, the        multiplexer being operable to provide the upper trip voltage and        the lower trip voltage time-sequentially as a trip voltage to        the non-inverting input of the comparator.

The comparator thus is operable to compare the converter current sensingvoltage to either the upper trip voltage or the lower trip voltage, tooutput the switching control voltage at the first logic level inresponse to each crossover of the lower trip voltage by the convertercurrent sensing voltage in the negative direction, and to output theswitching control voltage at the second logic level in response to eachcrossover of the upper trip voltage by the converter current sensingvoltage in the positive direction.

The outputting of the switching control voltage at the first logic levelis associated with the increase time duration. The outputting of theswitching control voltage at the second logic level is associated withthe decrease time duration.

In an embodiment, the switch-mode converter is arranged for charging anddischarging an inductive output filter, the inductive output filterbeing, during use, in electrical communication with the LED circuitarrangement.

The switch-mode converter is thus operable to control the increase ofthe converter current from the valley current level to the peak currentlevel in response to the switching control voltage equalling the firstlogic level over the increase time duration, and operable to control thedecrease of the converter current from the peak current level to thevalley current level in response to the switching control voltageequalling the second logic level over the decrease time duration.

The inductive output filter may be comprised in the switch-modeconverter, or alternatively be externally connected to the switch-modeconverter or the circuit arrangement.

-   -   The switch-mode converter may further comprise a component        selected from the group consisting of a diode and a second        switch,    -   the second switch being in electrical communication with the        hysteretic comparator to be closed and opened as a function of        the switching control voltage,    -   the component being in electrical communication with the switch        via an output node, and    -   the output node being, during use, in electrical communication        with the LED circuit arrangement.

With these components, a so-called half-bridge structure is constructedallowing to switch the output node between an upper and a lower supplyvoltage.

The switch-mode converter may comprise an inductive output filter, theinductive output filter being, during use, in electrical communicationwith the LED circuit arrangement.

The inductor may be changed and discharged, the charging and dischargingbeing controlled by the switch of the switch-mode converter. Chargingand discharging of the inductor may be associated with increasing anddecreasing of the converter current.

The inductor may, e.g. be connected to a half-bridge structure describedabove to form a so-called Buck converter, Buck-Boost converter or Boostconverter.

In an embodiment, the converter current sensor is arranged to determinethe converter current sensing voltage from a voltage drop over aresistor, the resistor being arranged to transmit the converter currentflowing through LED circuit arrangement.

The resistor can be outside or inside the circuit arrangement. Inparticular, when the circuit arrangement is integrated in an integratedcircuit, the resistor is preferably outside the circuit arrangement.

In a further embodiment, the circuit arrangement comprises the resistorand the resistor is in electrical communication with the LED circuitarrangement and with the converter current sensor.

In an embodiment of the circuit arrangement, the circuit arrangementfurther comprises:

-   -   a power supply operable to deliver an input supply voltage, the        power supply being in electrical communication with the        switch-mode converter to supply the switch-mode with the input        supply voltage, and    -   a capacitive input filter in electrical communication with the        power supply.

The capacitive input filter is usually applied to reduce sensitivity tovariations in the supply voltage, in particular to reduce thesensitivity to disturbances on the supply voltage. Usually withprior-art hysteretic control, a strong filtering is required with alarge capacitor, because at the lower conversion frequencies inputcurrent is drawn from the input capacitor for a relatively longduration. With the invention, a less strong filtering can be accepted,and a smaller capacitor can be applied, which may be attractive becauseof cost considerations.

In an embodiment of the circuit arrangement,

-   -   a LED controller is in electrical communication with the LED        circuit arrangement, and    -   the LED circuit arrangement comprises a first LED segment and at        least a second LED segment, the first LED segment being        associated with a first switching element operable for        controlling a path of the LED current through the first LED        segment, the second LED segment being associated with a second        switching element operable for controlling a path of the LED        current through the second LED segment,    -   the first and second switching elements being operable by the        LED segment controller for controlling a path of the LED current        flowing through the LED circuit arrangement.

The LED controller controls the path of the LED current flowing throughthe LED circuit arrangement by operating the first and the secondswitching element for controlling the path of the current through thefirst and the second LED segments is associated with varying the load ofthe LED circuit arrangement. The effects of this load variation may beeffectively diminished by controlling the converter current period.

In a further embodiment,

-   -   the first switching element is electrically parallel to the        first LED segment,    -   the second switching element is electrically parallel to the        second LED segment, and    -   the first and second switching elements are operable by the LED        segment controller for selecting the path of the LED current to        pass through the LED segment associated with the switching        element or to bypass the LED segment associated with the        switching element.

When the first switching element is open, the current will flow throughthe first LED segment. When the first switching element is closed, thecurrent will flow through the first switching element and bypass thefirst LED segment.

When the second switching element is open, the current will flow throughthe second LED segment. When the second switching element is closed, thecurrent will flow through the second switching element and bypass thesecond LED segment.

By operating the first and second switching elements, the path of theLED current is thus selected to pass selectively through the LEDsegments.

In an embodiment, the LED circuit load detector is in electricalcommunication with the LED controller and wherein the LED circuit loaddetector is arranged to cooperate with the LED controller to determinethe LED circuit load.

The LED circuit load may thus be directly obtained and used forcontrolling the converter current period, without the need for anadditional measurement on an electrical signal.

The LED driver IC according to the invention comprises one of thecircuit arrangements described above.

The LED driver IC may include one or more of the above-mentionedcomponents like inductors, capacitors and/or resistors, but thesecomponents may also be external to the IC, and connected to the ICduring use to cooperate with the IC. The composition if the externalcomponent and the IC may then together form a complete circuitarrangement according to the invention.

In further embodiments, the LED driver IC according to the inventioncomprises a plurality of any one or more of the circuit arrangementsdescribed above, each of the plurality of circuit arrangements beingassociated with a corresponding LED circuit arrangement.

The LED driver IC is thus operable to regulate a first current with afirst mean current level through a first LED circuit arrangement, e.g.comprising a red and an amber LED, and a second current with a secondmean current level through a second LED circuit arrangement, e.g.comprising a green and a blue LED.

The first and the second mean current level may be different, e.g. toaccommodate for the different physical structure of a first class of redand amber LEDs compared to a second class of green and blue LEDs.

The circuit arrangements in the plurality of any one or more of thecircuit arrangements may be of the same type, but may alternatively beof different types. E.g., a LED driver IC with two circuit arrangementsmay comprise a first and a second circuit arrangement, each comprising aswitch-mode converter according to a Buck topology. Alternatively, a LEDdriver IC with two circuit arrangements may, e.g. comprise a firstcircuit arrangement comprising a switch-mode converter according to aBuck topology and a second circuit arrangement comprising a switch-modeconverter according to a Buck-Boost topology.

The invention further provides a circuit composition comprising:

-   -   a first arrangement selected from the group consisting of a LED        circuit arrangement and a LED driver IC, as described above, and    -   a LED circuit arrangement including at least one LED, wherein        the first arrangement is in electrical communication with the        LED circuit arrangement for regulating a LED current flowing        through the LED circuit arrangement.

The circuit composition has the advantageous behaviour that the LEDcurrent is regulated with a well-controlled mean current level andwell-controlled converter current period.

In an embodiment of the circuit composition, the LED circuit arrangementcomprises a first LED segment and a second LED segment, the first LEDsegment being associated with a first switching element operable forcontrolling a path of the LED current through the first LED segment, thesecond LED segment being associated with a second switching elementoperable for controlling a path of the LED current through the secondLED segment,

-   -   the first and second switching elements being operable by the        LED segment controller for controlling a path of the LED current        flowing through the LED circuit arrangement.

In a further embodiment of the circuit composition,

-   -   the first switching element is electrically parallel to the        first LED segment,    -   the second switching element is electrically parallel to the        second LED segment, and    -   the first and second switching elements being operable by the        LED segment controller for selecting the path of the LED current        to pass through the LED segment associated with the switching        element or to bypass the LED segment associated with the        switching element.

A further embodiment of the invention relates to a LED lighting systemcomprising a LED circuit arrangement including at least one LED and oneof the circuit arrangements described above.

The LED lighting system may comprise any one of the circuit compositionsdescribed above.

The LED lighting system may be a brightness controlled LED-lamp, acolor-variable LED lamp, a LED matrix light source, a LED matrixdisplay, a large-sized LED information display for advertisement ormoving images, a LED-backlight for a LCD-TV, a LED-backlight for aLCD-monitor, or any other lighting system in which the current level ofthe LED current through at least one LED may be regulated in accordancewith aspects of the present invention as described above.

BRIEF DESCRIPTION OF DRAWINGS

The above and other aspects of the invention will be further elucidatedand described in detail with reference to the drawings, in whichcorresponding reference symbols indicate corresponding parts:

FIG. 1 a schematically shows a circuit arrangement according to theprior art, supplying a current to a fixed LED arrangement;

FIG. 1 b shows electrical signals related to the circuit arrangement ofFIG. 1 a;

FIG. 2 schematically shows the circuit arrangement according to theprior art, supplying a current to a switchable LED arrangement;

FIG. 3 schematically shows a block diagram of an embodiment of a circuitarrangement according to the invention;

FIGS. 4 a-4 c schematically show block diagrams of embodiments ofcircuit arrangements according to the invention;

FIGS. 5 a and 5 b show exemplary embodiments of a hysteretic comparatorusable in embodiments of a circuit arrangement according to theinvention;

FIG. 6 a and FIGS. 6 c-6 e show exemplary embodiments of switch-modeconverters of the Buck-converter type and the associated convertercurrent sensor usable in embodiments of a circuit arrangement accordingto the invention, electrically connected to an exemplary embodiment of aLED circuit arrangement; FIG. 6 b shows electrical signals related tothe embodiment of FIG. 6 a; FIG. 6 f shows electrical signals related tothe embodiment of FIG. 6 e;

FIGS. 7 a and 7 d show exemplary embodiments of switch-mode convertersof the Buck-Boost converter type, electrically connected to an exemplaryembodiment of a LED circuit arrangement; FIGS. 7 b and 7 c showselectrical signals related to the embodiment of FIG. 7 a without andwith an optional capacitor in the LED circuit arrangement;

FIGS. 8 a-8 b show exemplary embodiments of the converter current periodcontroller usable in embodiments of a circuit arrangement according tothe invention;

FIGS. 9 a-9 b show exemplary embodiments of the power supply usable inembodiments of a circuit arrangement according to the invention;

FIG. 10 shows a LED circuit arrangement with bypass switches;

FIG. 11 shows a simulation of the electrical characteristics as afunction of time of a circuit arrangement when operated with a LEDcircuit arrangement;

FIG. 12 shows a further embodiment of the converter current periodcontroller;

FIG. 13 shows another further embodiment of the converter current periodcontroller;

FIG. 14 shows an exemplary embodiment of a resistive digital-to-analogueconverter usable in embodiments of a circuit arrangement according tothe invention;

FIG. 15 a shows a circuit composition comprising a LED driver IC and aLED circuit arrangement according to the invention;

FIG. 15 b shows another circuit composition comprising a LED driver ICand a LED circuit arrangement according to the invention;

FIG. 16 shows an alternative circuit composition comprising a LED driverIC and a LED circuit arrangement according to the invention;

FIG. 17 shows an embodiment of a LED lighting system according to theinvention.

DETAILED DESCRIPTION

FIG. 1 a schematically shows a circuit arrangement CIRC according to theprior art, supplying a current to a fixed LED arrangement LEDCIRC. FIG.1 b shows electrical signals related to the circuit arrangement CIRCshown in FIG. 1 a.

The circuit is arranged for regulating a mean current level of a LEDcurrent ILED flowing through a LED arrangement LEDCIRC. In the exampleshown, the LED arrangement LEDCIRC is a series arrangement of a firstlight emitting diode LED1 and a second light emitting diode LED2. Theexample shown uses a so-called hysteretic Buck-converter, in which thefull converter current IL flowing through the circuit arrangement is fedto the LED circuit arrangement as the LED current ILED.

The circuit arrangement has a converter current sensor ILSEN. Theconverter current sensor includes a sense resistor RS, which is arrangedto conduct the converter current IS. The voltage drop over the senseresistor RS is representative of the current level of the convertercurrent IL. The voltage drop will be further referred to as theconverter current sensing voltage VS.

The circuit arrangement further comprises a hysteretic comparator HCOMPand a switch-mode converter SMCONV. The hysteretic comparator HCOMP canestablish an upper trip voltage VH and a lower trip voltage VL ascontrol crossover thresholds. The upper trip voltage VH is associatedwith a peak current level ILH of the converter current IL and the lowertrip voltage VL is associated with a valley current level ILL of theconverter current IL. The mean current level ILAVE is an average of thepeak current level ILH and the valley current level of the convertercurrent ILL. The hysteretic comparator HCOMP is in electricalcommunication with the converter current sensor ILSEN to receive theconverter current sensing voltage VS.

The converter current sensing voltage VS is connected to an invertinginput of a comparator CMP. The non-inverting input of the comparator CMPis connected via a multiplexer MUX to either the lower trip voltage VLor the upper trip voltage VH. Feedback from the output of the comparatorCMP to the multiplexer MUX selects either the lower trip voltage VL orthe upper trip voltage VH. In response to each crossover of the lowertrip voltage VL by the converter current sensing voltage VS in anegative direction, the comparator CMP and hysteretic comparator HCOMPthus output a switching control voltage VSW at a first logic LHL. Inresponse to each crossover of the upper trip voltage VH by the convertercurrent sensing voltage VS in a positive direction, the comparator CMPand the hysteretic comparator HCOMP output the switching control voltageVSW at a second logic level LLL.

The switch-mode converter SMCONV is operable to control a flow of theLED current ILED through the LED circuit arrangement LEDCIRC bycontrolling a flow of the converter current IL through the circuitarrangement CIRC. The switch-mode converter SMCONV is in electricalcommunication with the hysteretic comparator HCOMP to receive theswitching control voltage VSW.

In response to the switching control voltage VSW equalling the firstlogic level LHL, the switch-mode converter SMCONV controls an increaseof the converter current IL from the valley current level to the peakcurrent level. This controlling of the increase of the converter currentIL will continue for a time duration which will be further referred toas an increase time duration TH. In response to the switching controlvoltage VSW equalling the second logic level, the switch-mode converterSMCONV controls a decrease of the converter current IL from the peakcurrent level to the valley current level. This controlling of thedecrease of the converter current IL will continue for a time durationwhich will be further referred to as a decrease time duration TL.

The circuit arrangement CIRC will thus supply the LED circuitarrangement LEDCIRC with a LED current at the mean current level. TheLED current oscillates with a converter current period T between avalley current level and a peak current level. The converter currentperiod T comprises the increase time duration TH and the decrease timeduration TL. The valley current level and the peak current level aredependent on the upper trip voltage VH and the lower trip voltage VLrespectively. The difference between the peak current level and thevalley current level will be further referred to as a peak-to-peakcurrent ripple dI. The mean current level is dependent on the meanvoltage level of the upper trip voltage VH and the lower trip voltageVL, referred to as a reference voltage level VREF. The differencebetween the upper trip voltage VH and the lower trip voltage VL will bereferred to as the hysteresis voltage VHYS. The increase time durationTH, the decrease time duration TL and hence also the converter currentperiod T, depend on these voltages and may be further dependent on, e.g.the circuit load of the LED circuit.

The switch-mode converter comprises a switch SW1, a diode D2 and aninductor L1. The inductor L1 is connected between an intermediate nodeLX, located in between the switch SW1 and the diode D2, and the LEDcircuit arrangement LEDCIRC. The switch SW1 and the diode D2 switch theLX node to an input voltage Vin, supplied by an external DC powersupply, or to ground GND, depending on the state of the switch SW1.Switching the LX node to the input voltage Vin or to ground GNDrespectively charges and discharges the inductor L1 and consequentlyincreases or decreases the current level of the converter current IL.

This class of circuit arrangements CIRC will be referred to as examplesof hysteretic converters. The specific switch-mode converter describedhere will be referred to as an example of a converter according to aso-called Buck-converter topology. Alternative converter topologies mayalso be feasible within the scope of the invention. As an example, alsoan alternative hysteretic converter, such as a so-called hystereticBuck-Boost-converter, or a so-called hysteretic Boost-converter may beused in embodiments of the invention.

FIG. 2 schematically shows the circuit arrangement CIRC according to theprior art, supplying a current to a switchable LED circuit arrangementLEDCIRC.

The circuit arrangement CIRC may be the same as described in referencewith FIG. 1. The LED circuit arrangement LEDCIRC however comprises afirst LED LED1 and a second LED LED2, each associated with a respectiveswitching element B1, B2. The first switching element B1 is electricallyparallel to the first LED LED1 and the second switching element B2 iselectrically parallel to the second LED LED2. The first and secondswitching elements B1, B2 are each operable by a LED segment controllerPWMCON for selecting the path of the LED current to pass through the LEDassociated with the respective switching element or to bypass the LEDassociated with the respective switching element. The LED arrangementthus allows to vary the effective light output of each of the two LEDsindividually, by varying the time that the path of the LED currentpasses through a LED with a duty cycle of a control period. The controlperiod is generally a period of a fixed length, also referred to as apulse width modulation period corresponding to a pulse width modulationfrequency. The duty cycle associated with operating the first LED LED1to emit light is further referred to as the first LED duty cycle PWM1.The duty cycle associated with operating the second LED LED2 to emitlight is further referred to as the second LED duty cycle PWM2.

It should be noted that in stead of a single LED, also, e.g. a pluralityof LEDs arranged in series may be used and operated by a singleswitching element in parallel to the series arrangement of the pluralityof LEDs. This may also be referred to as a LED segment. The LEDs fromthe plurality of LEDs in a single LED segment may have substantially thesame colour, but the colours may also be different between the LEDswithin a segment. When referring to “LED” in the following, it shall beunderstood to also refer to embodiments using a “LED segment” comprisinga plurality of LEDs.

When the two LEDs LED1, LED2 are operated with duty cycles PWM1, PWM2using the switches B1, B2, the circuit load of the LED circuitarrangement will differ as a consequence of the different voltage dropVLED over the LED circuit arrangement depending on which switch is openand which is closed. The hysteretic converter will however maintain themean current level as the same value, due to the operating mechanismdescribed above in reference with FIG. 1. However, in doing so, the LEDcurrent frequency—or the LED current period or the, for this convertertype equivalent, converter current period—of the oscillation of the LEDcurrent will differ.

For the circuit arrangement shown, the mean current of the LED currentILED may be described in terms of the upper trip voltage VH, the lowertrip voltage VL and the sense resistor value RS as:ILED=(VH+VL)/(2·RS)  (1)

The peak-to-peak ripple dI of the LED current is given by the differencebetween the upper trip voltage VH and the lower trip voltage VL and thesense resistor value RS:dI=(VH−VL)/RS  (2)

The charge and discharge rate dI/dt_(c) and dI/dt_(d) of the inductor L1with an inductive value L is determined by the supply voltage Vin andthe load voltage VLED in the following way:Charging: dI/dt _(c)=(Vin−VLED)/L  (3)Discharging: dI/dt _(d) =−VLED/L  (4)Consequently, the LED current control period T, being the sum of thecharging time t_(c) and the discharging time t_(d), or the sum of theincrease time duration TH and the decrease time duration TL, is givenby:T=TH+TL=t _(c) +t _(d)=L·dI/(Vin−VLED)+L·dI/Vled=L·(VH−VL)/(RS·(Vin−VLED))+L·(VH−VL)/(RS·VLED)  (5)

From (5) it can be seen that the LED current frequency f=1/T variesconsiderably with varying load voltage VLED, having its maximum atVLED=Vin/2 and its minima at VLED close to 0 and VLED close to Vin(determined by possible further resistances in the charge and dischargepath, which are ignored in the formulas above).

In this example with two bypass switches four states will occur for theload voltage VLED. Expressing the voltage over the first LED1 with afirst LED voltage VLED1 and the voltage over the second LED2 with asecond voltage VLED2, and ignoring the voltage drop over a closed(conducting) switch B1, B2, the four load voltages are:both switches B1, B2 off: VLED=VS+VLED1+VLED2;only switch B1 on; switch B2 off: VLED=VS+VLED2;only switch B2 on; switch B2 off: VLED=VS+VLED1;both switches B1, B2 on: VLED=VS.

This results in four different frequencies when using a circuitarrangement CIRC according to the prior art. Of these frequencies, threefrequencies are associated with conditions during which at least one LEDemits light. For all four frequencies, the mean current level issubstantially constant.

FIG. 3 schematically shows a block diagram of an embodiment of a circuitarrangement CIRC according to the invention, in electrical communicationwith a led circuit arrangement LEDCIRC. Details of the differentelements of the circuit arrangement CIRC and the LED circuit arrangementLEDCIRC are not drawn, but will be described for different embodimentsfurther below.

In FIG. 3, the circuit arrangement comprises a converter current sensorILSEN, a hysteretic comparator HCOMP, a switch-mode converter SMCONV, aconverter current period controller LPCON and a power supply VINGEN.

The converter current sensor ILSEN may be of the type described inreference with FIG. 1, and is in electrical communication with at leastthe hysteretic comparator HCOMP, the switch-mode converter SMCONV andthe converter current period controller LPCON.

The switch-mode converter SMCONV may be of the type described inreference with FIG. 1, and is in electrical communication with at leastthe converter current sensor ILSEN to receive the converter current IL,the hysteretic comparator HCOMP to receive a switching control voltageVSW and, the power supply VINGEN to receive the input voltage Vin.

The hysteretic comparator HCOMP is in electrical communication with theswitch-mode converter SMCONV and the converter current period controllerLPCON. The hysteretic comparator HCOMP is arranged to receive a firsttrip control voltage VC1 and a second trip control voltage VC2 from theconverter current period controller LPCON. The hysteretic comparatorHCOMP is operable to establish the upper trip voltage VH and the lowertrip voltage VL from the first trip control voltage VC1 and the secondtrip control voltage VC2. The hysteretic comparator HCOMP may furtheroperate like the hysteretic comparator HCOMP described in reference withFIG. 1, and is thus operable to output a switching control voltage VSWto the switch-mode converter SMCONV.

The converter current period controller LPCON is a new and inventiveelement to the circuit arrangement according to the invention. Theconverter current period controller LPCON is operable to control aconverter current period T to be within a period control range Tref.

The period control range Tref may be defined with a lower periodthreshold TrefL (not shown) and an upper period threshold TrefH (notshown). The converter current period T may then be controlled to bebetween the lower period threshold TrefL and the upper period thresholdTrefH.

The lower period threshold TrefL and the upper period threshold TrefHmay be determined from a centre duration and a duration width. Theconverter current period T may then be controlled to be within theduration width around the centre duration. The duration width ispreferably smaller than 10% of the centre duration, such that theconverter current period T varies with only a small fraction, i.e. withless than 10%. In an example, the centre duration corresponds to theperiod of a frequency of 1 MHz and the duration width is 100 kHz,resulting in a lower period threshold TrefL associated with a frequencyof 0.95 MHz and an upper period threshold TrefH associated with afrequency of 1.05 MHz. In this example, the centre duration has a fixedvalue, but the centre duration may also be varied real-time over aspectral band.

The converter current period controller LPCON may comprise a convertercurrent period detector FDET operable to establish an indicator INDassociated with the converter current period T. The indicator IND may bea measurement value of the converter current period, an estimate of theconverter current period, or any other indicator that can be used toestablish a suitable first and second trip control voltage forcontrolling the converter current period.

In establishing the indicator IND, a plurality of indicator values maybe acquired and filtered in order to obtain the indicator IND. Thefiltering may comprise a low-pass filtering in order to reduce thesensitivity to high-frequent fluctuations, e.g. due to noise or effectsof quantization, on the acquired indicator values.

As shown in FIG. 4 a, the converter current period detector FDET may bein electrical communication with the converter current sensor ILSEN toreceive the converter current sensing voltage VS. The converter currentperiod detector FDET may measure a value of the converter current periodfrom the current sensing voltage VS, and establish the indicator INDfrom this measurement. A suitable first trip control voltage VC1 andsecond trip control voltage VC2 are then derived from this indicatorIND.

As shown in FIG. 4 b, the converter current period detector FDET may bein electrical communication with the hysteretic comparator HCOMP toreceive the switching control voltage VSW. The converter current perioddetector FDET may measure a value of the converter current period fromthe switching control voltage VSW, and establish the indicator IND fromthis measurement. A suitable first VC1 and second trip control voltageVC2 are then derived from this indicator IND.

As shown in FIG. 4 c, the converter current period detector FDET may bein electrical communication with a LED circuit load detector (LDET) toreceive a LED circuit load. The LED circuit load detector (LDET) may bein electrical communication with the LED circuit arrangement LEDCIRC.The converter current period detector FDET may estimate a value of theconverter current period T from the LED circuit load, and establish theindicator IND from this estimation. A suitable first VC1 and second tripcontrol voltage VC2 are then derived from this indicator IND. Theconverter current period detector FDET may alternatively directly usethe LED circuit load as the indicator IND, and use the indicator INDe.g. for a look-up in a look-up table in order to retrieve a suitablefirst VC1 and second trip control voltage VC2. E.g., when the LEDcircuit load is expressed as the status of the switching elements B1,B2, the status of the switching elements B1, B2 can be used forretrieval from the look-up table.

FIGS. 5 a and 5 b show exemplary embodiments of the hystereticcomparator HCOMP being a part of an embodiment of a circuit arrangementCIRC according to the invention. The hysteretic comparator HCOMP isoperable to receive the first and second trip control voltages VC1, VC2from the converter current period detector FDET in a voltageestablishing unit VEST. The voltage establishing unit VEST is operableto establish the upper and lower trip voltages VH, VL in response to thefirst and second trip control voltages VC1, VC2.

The first and second trip control voltages VC1, VC2 may be the uppertrip voltage and lower trip voltage required to be applied forcontrolling the converter current period to be within the period controlrange Tref. The hysteretic comparator HCOMP may then use the first andsecond trip control voltages VC1, VC2 as the new upper and lower tripvoltages VH, VL.

Alternatively, the first and second trip control voltages VC1, VC2 maybe adjustment values to the upper and lower trip voltages VH, VL asbeing applied. The hysteretic comparator HCOMP may then add the firstand second trip control voltages VC1, VC2 to the upper and lower tripvoltages VH, VL to obtain new upper and lower trip voltages VH, VL.

In a first embodiment shown in FIG. 5 a, the non-inverting input of afirst comparator CMP1 is connected to the upper trip voltage VH and thenon-inverting input of a second comparator CMP2 is connected to thelower trip voltage VL or the upper trip voltage VH. The convertercurrent sensing voltage VS is connected to the inverting input of thefirst comparator CMP1 and to the inverting input of the secondcomparator CMP2. The output of the first comparator CMP1 and the outputof the second comparator CMP2 are fed into a digital multiplexer MUXD.Feedback from the output of the digital multiplexer MUXD to the digitalmultiplexer MUXD selects output of the first comparator CMP1 and theoutput of the second comparator CMP2 as the switching control voltageVSW on the output of the digital multiplexer MUXD. It will be clear to aperson skilled in the art that the selection performed by the digitalmultiplexer MUXD may alternatively be implemented using alternativeelectrical components, e.g. a flip-flop. In response to each crossoverof the lower trip voltage VL by the converter current sensing voltage VSin a negative direction, the hysteretic comparator HCOMP thus outputsthe switching control voltage VSW at a first logic LHL. In response toeach crossover of the upper trip voltage VH by the converter currentsensing voltage VS in a positive direction, the hysteretic comparatorHCOMP output the switching control voltage VSW at a second logic levelLLL.

In a second embodiment shown in FIG. 5 b, the non-inverting input of acomparator CMP is connected via a multiplexer MUX to either the lowertrip voltage VL or the upper trip voltage VH. This hysteretic comparatorconstruction has the advantage first that there is only one comparatoroffset error contribution, whereas the construction of FIG. 5 a has twolikely different comparator offset error contributions from each of thetwo comparators. Moreover, the comparator offset is in the samedirection both for VH and VL yielding an inherent accurate hysteresewindow VH−VL, as the comparator offset effectively cancels out in thedifference VH−VL. The converter current sensing voltage VS is connectedto an inverting input of the comparator CMP. Feedback from the output ofthe comparator CMP to the multiplexer MUX selects either the lower tripvoltage VL or the upper trip voltage VH as a trip voltage VTR on theoutput of the multiplexer MUX. In response to each crossover of thelower trip voltage VL by the converter current sensing voltage VS in anegative direction, the comparator CMP and hysteretic comparator HCOMPoutput a switching control voltage VSW at a first logic LHL. In responseto each crossover of the upper trip voltage VH by the converter currentsensing voltage VS in a positive direction, the comparator CMP and thehysteretic comparator HCOMP output the switching control voltage VSW ata second logic level LLL.

FIGS. 6 a-6 d show exemplary embodiments of switch-mode converters ofthe Buck-converter type and the associated converter current sensor asused in embodiments of a circuit arrangement according to the invention.FIGS. 6 a-6 d further show exemplary and non-limiting embodiments of aLED circuit arrangement fed by the switch-mode converter.

The switch-mode converter SMCONV shown in FIG. 6 a includes a switchSW1, a diode D2 and an inductor L1. The switch-mode converter SMCONV issimilar to the one described in reference to FIG. 1. The inductor L1 isconnected between an intermediate node LX, located in between the switchSW1 and the diode D2, and the LED circuit arrangement LEDCIRC. Theswitch SW1 and the diode D2 switch the LX node to an input voltage Vin,supplied by an external DC power supply, or to ground GND, depending onthe state of the switch SW1. Switching the LX node to the input voltageVin or to ground GND respectively charges and discharges the inductor L1and consequently increases or decreases the current level of theconverter current IL.

FIG. 6 b shows electrical signals related to the embodiment of FIG. 6 a.Curve cL1 shows the converter current IL as a function of time, i.e. thecurrent through the inductor L1 and through the sense resistor RS. CurvecVS shows the current sensing voltage VS as a function of time. CurvecILED shows the LED current ILED as a function of time, i.e. the currentflowing through the LEDs (when the associated bypass switch in the LEDcircuit arrangement is open). Curve cSW1 shows the switching controlvoltage VSW, which can take a high logic level LHL corresponding to aclosed switch SW1 or a low logic level LLL corresponding to an openswitch SW1. Curve cLX shows the voltage at node LX, which can take a lowvalue corresponding to ground or a high value corresponding to the inputvoltage Vin.

The switch-mode converter SMCONV is operable to control a flow of theLED current ILED with a mean current level through the LED circuitarrangement LEDCIRC. The switch-mode converter SMCONV is in electricalcommunication with the hysteretic comparator HCOMP to receive theswitching control voltage VSW.

In response to the switching control voltage VSW equaling the firstlogic level LHL, the switch-mode converter SMCONV controls an increaseof the converter current IL from the valley current level to the peakcurrent level, as is shown in FIG. 6 b. This controlling of the increaseof the converter current will continue for a time duration which will befurther referred to as an increase time duration TH. This phase ofcontrolling may be further referred to as a increase phase pH. Inresponse to the switching control voltage VSW equaling the second logiclevel, the switch-mode converter SMCONV controls a decrease of theconverter current IL from the peak current level to the valley currentlevel. This controlling of the decrease of the converter current willcontinue for a time duration which will be further referred to as adecrease time duration TL. This phase of controlling may be furtherreferred to as a decrease phase pL.

The circuit arrangement CIRC will thus supply the LED circuitarrangement LEDCIRC with a LED current at the mean current level, whichoscillates with a converter current period T between a valley currentlevel and a peak current level. The valley current level and the peakcurrent level are dependent on the upper trip voltage VH and the lowertrip voltage VL respectively. The difference between the peak currentlevel and the valley current level will be further referred to as apeak-to-peak current ripple dI. The mean current level is dependent onthe mean voltage level of the upper trip voltage VH and the lower tripvoltage VL, referred to as a reference voltage level VREF. Thedifference between the upper trip voltage VH and the lower trip voltageVL will be referred to as the hysteresis voltage VHYS. The increase timeduration TH, the decrease time duration TL and hence also the convertercurrent period T, depend on these voltages and may be further dependenton, e.g. the circuit load of the LED circuit.

The converter current sensor ILSEN shown in FIG. 6 a comprises a senseresistor RS in the current path of the converter current IL and isoperable to perform a voltage measurement over the sense resistor RS.The measured voltage may be outputted as the converter current sensingvoltage VS.

The switch-mode converter SMCONV shown in FIG. 6 c includes a switchSW1, a second switch SW2 and an inductor L1. The inductor L1 isconnected between an intermediate node LX, located in between the switchSW1 and the second switch SW2, and the LED circuit arrangement LEDCIRC.The switch SW1 and the second switch SW2 switch the LX node to an inputvoltage Vin, supplied by an external DC power supply, or to ground GND,depending on the state of the switch SW1 and SW2. Switching the LX nodeto the input voltage Vin or to ground GND respectively charges anddischarges the inductor L1 and consequently increases or decreases thecurrent level of the converter current IL.

The switch-mode converter SMCONV is in electrical communication with thehysteretic comparator HCOMP to receive the switching control voltage VSWvia a break-before-make circuit BBM. The break-before-make circuit BBMcomprises a timing circuit which assures that the two switches SW1 andSW2 can not both be closed at the same time, as this would result in ashort circuit of the input voltage Vin and the ground voltage. Thebreak-before-make circuit BBM is thus operable to generate a first and asecond switching control voltage VSW1, VSW2 from the switching controlvoltage VSW which operate the switches SW1 and SW2 to never be closedsimultaneously.

The converter current sensor ILSEL in FIG. 6 c is similar to the oneshown in FIG. 6 a.

Electrical signals related to the embodiment of FIG. 6 c aresubstantially the same as the electrical signals shown in FIG. 6 b inrelation with the embodiment of FIG. 6 a, and are not drawn again.

FIG. 6 d shows the same switch-mode converter SMCONV as FIG. 6 c and analternative converter current sensor ILSEN.

The sense resistor RS is placed in the current path of the convertercurrent IL, outside the converter current sensor ILSEN. The convertercurrent sensor ILSEN is operable to perform a voltage measurement overthe sense resistor RS which is external to the converter current sensorILSEN. The measured voltage may be outputted as the converter currentsensing voltage VS.

An advantage of this embodiment of the converter current sensor ILSEN isthat it allows integration of the converter current sensor ILSEN in anintegrated circuit, such as a LED driver IC. The value of the senseresistor RS may then be selected independently of the LED driver IC,allowing an wider possibility of selecting the current level that theLED driver IC can support.

Electrical signals related to the embodiment of FIG. 6 d aresubstantially the same as the electrical signals shown in FIG. 6 b inrelation with the embodiment of FIG. 6 a, and are not drawn again.

FIG. 6 e shows a similar switch-mode converter SMCONV as FIG. 6 dcooperating with the same alternative converter current sensor ILSEN asin FIG. 6 c.

In FIG. 6 e, the LED circuit arrangement LEDCIRC further comprises ancapacitive filter C1, connected in parallel to the series circuit of theLED circuit arrangement.

FIG. 6 f shows electrical signals related to the embodiment of FIG. 6 e.Curve cL1C shows the converter current IL as a function of time, i.e.the current through the inductor L1 and through the sense resistor RS.Curve cVSC shows the current sensing voltage VS as a function of time.Curve cILEDC shows the LED current as a function of time, i.e. thecurrent flowing through the LEDs (when the associated bypass switch inthe LED circuit arrangement is open). Curve cSW1C shows the switchingcontrol voltage VSW, which can take a high logic level LHL correspondingto a closed switch SW1 or a low logic level LLL corresponding to an openswitch SW1. Curve cLXC shows the voltage at node LX, which can take alow value corresponding to ground or a high value corresponding to theinput voltage Vin.

Comparing the curves in FIG. 6 f with the curves in FIG. 6 b, it can beobserved that the capacitive filter C1 provides a smoothing of the LEDcurrent amplitude, with the beneficial effects that the lifetime of theLEDs is increased since the peak current level through the LEDs isreduced. Also the ripple amplitude of the LED current is reduced,reducing the ripple amplitude of the light level. Alternatively, with acapacitive filter C1, a larger fluctuation of the converter current canbe allowed through the inductor to achieve the same LED current rippleamplitude as for a LED circuit arrangement without a capacitive outputfilter, which has the advantage of a smaller inductance value and size.

As a Buck-converter type switch-mode converter is not suitable for usewith a LED circuit arrangement LEDCIRC which may have a voltage dropVLED over the LED circuit arrangement LEDCIRC that is larger than theinput voltage Vin, a Buck-Boost converter topology may be preferred insome situations. The invention can also be applied with switch-modeconverters according to a Buck-Boost converter topology.

A first example of such a Buck-Boost switch-mode converters is shown inFIG. 7 a.

The switch-mode converter SMCONV shown in FIG. 7 a includes a switchSW1, a diode D2 and an inductor L1. The switch-mode converter SMCONV isconnected to current sense resistor RS of a converter current sensorILSEN. The inductor L1 is connected to the input voltage Vin via thecurrent sense resistor RS and an intermediate node LY. The inductor L1via an intermediate node LX to the switch SW1 which can connect toground GND. The LED circuit arrangement LEDCIRC is connected to theintermediate node LX via an intermediate node LZ and the diode D2, tothe input voltage Vin via a node LY. and to the inductor L1 via node LYand the sense resistor RS of the converter current sensor ILSEN.

An exemplary LED circuit arrangement LEDCIRC is shown with two LEDs in aseries arrangement. An optional capacitor C1 may be placed as ancapacitive filter between the input and output of the LED circuitarrangement, i.e. in parallel to the series arrangement of the LEDs, toprovide a smoothing of the LED current amplitude.

FIGS. 7 b and 7 c shows electrical signals related to the embodiment ofFIG. 7 a without and with an optional capacitor in the LED circuitarrangement respectively. Curves cL1BB and cL1BBC show the convertercurrent IL as a function of time, i.e. the current through the inductorL1 and through the sense resistor RS, for the LED circuit arrangementLEDCIRC without and with the capacitor C1 respectively. Curves cVSBB andcVSBBC show the current sensing voltage VS as a function of time. CurvecITRBB and cITRBBC show a transfer current ITR as a function of time,i.e. the current fed from the circuit arrangement CIRC to the LEDcircuit arrangement LEDCIRC. Curve cILEDBB and cILEDBBC show the LEDcurrent as a function of time, i.e. the current flowing through the LEDs(when the associated bypass switch in the LED circuit arrangement isopen). Curve cSW1BB and cSW1BBC show the switching control voltage VSW,which can take a high logic level LHL corresponding to a closed switchSW1 or a low logic level LLL corresponding to an open switch SW1. CurvescLXBB and cLXBBC show the voltage at node LX, which can take a low valuecorresponding to ground or a high value corresponding to the outputvoltage Vout.

To maintain Volt-second balance for the inductor L1 it can be seen thatthe output voltage Vout is always larger than the input voltage Vin.Since the LED circuit arrangement LEDCIRC is connected between Vout andVin, a voltage can be generated over the LED circuit arrangement LEDCIRCthat is smaller or larger than Vin, thus allowing to handle a wide rangeof load variation of the LED circuit arrangement.

Intermediate node LX switches to an output voltage Vout, or to groundGND, depending on the state of the switch SW1, as is shown by the curvecLXBB showing the voltage at node LX and the curve cSW1BB showing theswitching voltage VSW in FIG. 7 b and the curves cLXBBC and cSW1BBC inFIG. 7 c. Switching the LX node to the output voltage Vout or to groundGND respectively discharges and charges the inductor L1 and consequentlyincreases or decreases the current level of the converter current IL ina decrease phase pLBB, P1BBC and an increase phase pHBB, pHBBCrespectively.

In this example of the Buck-Boost converter feeding a LED circuitarrangement of a series arrangement of LEDs, a flow of the convertercurrent IL to the LED circuit arrangement, indicated in the Figures as atransfer current ITR, is prevented during the increase phase pHBB,pHBBC, in which the switch SW1 is closed, connecting node LX to groundGND, as is shown by curves cILEDBB in FIG. 7 b. The transfer current ITRis thus zero during the increase phase pHBB, pHBBC, and hence the LEDcurrent ILED is also zero when the LED circuit arrangement has nocapacitor C1. During the decrease phase pLBB, pLBBC, the switch SW2 isopen, the inductor L1 discharges via the diode D2 and the inductor L1thus feeds the converter current IL as the transfer current ITR to theLED circuit arrangement, such that the LED current is equal to theconverter current during the decrease phase pLBB, pLBBC. The mean LEDcurrent level thus is a weighted average of the peak current level andthe valley current level of the converter current.

For this exemplary embodiment using a Buck-Boost converter and denotingthe mean LED current level with ILEDave, the peak current level of theconverter current with ILH, the valley current level of the convertercurrent with ILL, the weighting average can be further expressed usingthe increase time duration TH, the decrease time duration TL, theconverter current period T, with T=TH+TL, as:ILEDave=(TL·(ILH+ILL)/2)/(TH+TL)  (6)

Expressing the peak current level and the valley current level of theconverter current in terms of the upper trip voltage VH and the lowertrip voltage VL, this may be rewritten as:ILEDave=(TL·(VH+VL)/(2·RS))/(TH+TL)=(TL/T)·(VH+VL)/(2·RS)  (7)

Alternatively, the weighting can be expressed using the output voltageVout, corresponding to the voltage over the series arrangement of LEDsin the LED circuit arrangement when all LEDs are emitting light, and theinput voltage Vin, asILEDave=(Vin/(Vin+Vout))·(VH+VL)/(2·RS)  (8)

An advantage of using the input and output voltage to obtain the weightsfor determining the mean LED current level is that no time durationsneed to be measured. Also, the output voltage Vout may also be used asthe indicator for the load.

When the optional capacitor C1 is present in the LED circuitarrangement, the transfer current ITR current feeding the LED circuitarrangement still has the same shape as curve cITRBB in FIG. 7 b, butthe LED current ILED flowing through (or bypassing) the LEDs is smoothedand behaves as shown as cILEDBBC in FIG. 7 c. Comparing the curves inFIG. 7 c with the curves in FIG. 7 b, it can be observed that thecapacitive filter C1 provides a smoothing of the LED current amplitude,with the beneficial effects that the lifetime of the LEDs is increasedsince the peak current level through the LEDs is reduced. Also theripple amplitude of the LED current is reduced, reducing the rippleamplitude of the light level. Alternatively, with a capacitive filterC1, a larger fluctuation of the converter current can be allowed throughthe inductor to achieve the same LED current ripple amplitude as for aLED circuit arrangement without a capacitive output filter, which hasthe advantage of a smaller inductance value and size.

The relation between the converter current, shown as cL1BB and cL1BBC inFIG. 7 b and FIG. 7 c respectively, and the LED current, shown as cLEDBBand cLEDBBC in FIG. 7 b and FIG. 7 c respectively, is taken into accountwhen determining the upper trip current level or upper trip voltagelevel and determining the lower trip current level or lower trip voltagelevel.

The converter current sensor ILSEL in FIG. 7 a is similar to the oneshown in FIG. 6 a, but may also be of a similar type as the one shown inFIG. 6 c.

An alternative switch-mode converter SMCONV shown in FIG. 7 b includes aswitch SW1, a second switch SW2 and an inductor L1.

The switch-mode converter SMCONV of FIG. 7 b is the same as that of FIG.7 a, but for the presence of a second switch SW2 replacing the diode D2,and the addition of a break-before-make circuit BBM.

The switch-mode converter SMCONV is in electrical communication with thehysteretic comparator HCOMP to receive the switching control voltage VSWvia the break-before-make circuit BBM. The break-before-make circuit BBMcomprises a timing circuit which assures that the two switches SW1 andSW2 can not both be closed at the same time, as this would result in ashort circuit of the output voltage Vout and the ground voltage. Thebreak-before-make circuit BBM is thus operable to generate a first and asecond switching control voltage VSW1, VSW2 from the switching controlvoltage VSW which operate the switches SW1 and SW2 to never be closedsimultaneously.

The converter current sensor ILSEL in FIG. 7 b is similar to the oneshown in FIG. 6 a, but may also be of a similar type as the one shown inFIG. 6 c.

FIG. 8 a shows a first exemplary embodiments of the converter currentperiod controller LPCON usable in embodiments of a circuit arrangementaccording to the invention. The converter current period controllerLPCON comprises a trip control voltage generator VCGEN operable toestablish a first trip control voltage VC1 and a second trip controlvoltage VC2.

The first and second trip control voltages VC1, VC2 may be voltages witha first and second trip control voltage level, and the trip controlvoltage generator VCGEN may be operable to establish these voltages tomake these voltages available to the hysteretic comparator HCOMP. Thevoltage establishing unit VEST in the hysteretic comparator HCOMP isoperable to establish the upper and lower trip voltages VH, VL inresponse to the first and second trip control voltages VC1, VC2 by, e.g.a direct electrical connection.

The first and second trip control voltages VC1, VC2 may alternatively abe first and second trip control voltage level value, and the tripcontrol voltage generator VCGEN may be operable to supply these voltagelevel values, e.g. as digital signals or digital register values, to thehysteretic comparator HCOMP. The voltage establishing unit VEST in thehysteretic comparator HCOMP is operable to establish the upper and lowertrip voltages VH, VL in response to the first and second trip controlvoltages VC1, VC2 in the form of trip control voltage level values by,e.g. a voltage generator.

FIG. 8 b shows a second exemplary embodiments of the converter currentperiod controller LPCON usable in embodiments of a circuit arrangementaccording to the invention. The converter current period controllerLPCON comprises a trip control voltage generator VCGEN operable toestablish a first trip control voltage VC1 and a second trip controlvoltage VC2 in cooperation with a voltage mixer VM.

In this example, the trip control voltage generator VCGEN is operable toestablish the reference voltage VREF and the hysteresis voltage VHYS,corresponding to the mean of the first and second trip control voltagesVC1, VC2 associated with the mean current level and the differencebetween the first and average control voltages VC1, VC2 associated withthe current ripple dI.

In this example, the voltage mixer VM is operable to determine the firstand second trip control voltage VC1, VC2 by adding half the hysteresisvoltage VHYS to the reference voltage VREF to achieve the first tripcontrol voltage VC1 and by subtracting half the hysteresis voltage VHYSfrom the reference voltage VREF to achieve the second trip controlvoltage VC2, as:VC1=VREF+VHYS/2,VC2=VREF−VHYS/2.

FIG. 9 a shows a first exemplary embodiment of the power supply VINGENusable in embodiments of a circuit arrangement according to theinvention. The power supply VINGEN is in electrical communication withthe switch-mode converter SMCONV. The power supply VINGEN is operable todeliver an input supply voltage Vin to the switch-mode converter SMCONV.

FIG. 9 b shows a second exemplary embodiment of the power supply VINGENusable in embodiments of a circuit arrangement according to theinvention. The power supply VINGEN is in electrical communication withthe switch-mode converter SMCONV. The power supply VINGEN is operable todeliver an input supply voltage Vin to the switch-mode converter SMCONV.A capacitive input filter Cin is in electrical communication with thepower supply VINGEN and with the switch-mode converter SMCONV, in orderto stabilize the input supply voltage Vin as received by the switch-modeconverter SMCONV.

FIG. 10 shows an example of a LED circuit arrangement LEDCIRC. The LEDcircuit arrangement LEDCIRC comprises a first LED LED1 and a second LEDLED2, each associated with a respective switching element B1, B2. Inthis example, the first LED LED1 is a series arrangement of three greenLEDs and the second LED LED2 is a series arrangement of two blue LEDs.

In this example, the voltage over the series arrangement of the threegreen LEDs is typically 10.8 V when a LED current with a mean currentlevel of approx. 700 mA is flowing through the green LEDs. The voltageover the series arrangement of the two blue LED is typically 7.2 V whena LED current with a mean current level of approx. 700 mA is flowingthrough the blue LEDs.

The first switching element B1 is electrically parallel to the first LEDLED1 and the second switching element B2 is electrically parallel to thesecond LED LED2. The first and second switching elements B1, B2 areoperable by a LED segment controller PWMCON for selecting the path ofthe LED current to pass through the LED associated with the switchingelement or to bypass the LED associated with the switching element. TheLED arrangement thus allows to vary the effective light output of eachof the two LEDs individually, by varying the time that the path of theLED current passes through a LED with a duty cycle of a control period.The relative ratio of light emitted by the green LED LED1 and blue LEDLED2 may thus be controlled. The control period is a period of a fixedlength, also referred to as a pulse width modulation periodcorresponding to a pulse width modulation frequency of 300 Hz. The dutycycle associated with operating the first LED LED1 to emit light isfurther referred to as the first LED duty cycle PWM1. The duty cycleassociated with operating the second LED LED2 to emit light is furtherreferred to as the second LED duty cycle PWM2.

In this example, the load corresponding to the LED circuit arrangementmay this take four typical values, dependent on the status of the firstand second switching elements B1, B2. In this example, the voltage overthe LED circuit arrangement may thus be typically 18.0 V when bothswitching elements B1, B2 are open and the LED current with a meancurrent level of approx. 700 mA is flowing through the green and theLEDs. The voltage over the LED circuit arrangement may thus be typically10.8 V (plus a small voltage drop over the switching element B2, whichis neglected in the discussion for simplicity) when switching element B1is open and switching element B2 is closed, such that the LED currentwith a mean current level of approx. 700 mA is flowing through the greenLEDs but bypassing the blue LEDs. Likewise, the voltage over the LEDcircuit arrangement may thus be typically 7.2 V when switching elementB1 is closed and switching element B2 is open, such that the LED currentwith a mean current level of approx. 700 mA is flowing through the blueLEDs but bypassing the green LEDs. Likewise, the voltage over the LEDcircuit arrangement may thus be close to 0 (only a small voltage dropover the switching elements B1, B2, which is neglected in the discussionfor simplicity) when switching element B1 and B2 are both closed, suchthat the LED current with a mean current level of approx. 700 mA isbypassing both the green LEDs and the blue LEDs.

FIG. 11 shows a simulation of the electrical characteristics as afunction of time of the circuit arrangement CIRC when operated with theLED circuit arrangement LEDCIRC shown in FIG. 10.

A first curve cPWM1 shows a control signal associated with the first LEDduty cycle PWM1 associated with the green LED LED1 and associated withoperation of the first switching element B1. A second curve cPWM2 showsa control signal associated with the second LED duty cycle PWM2associated with the blue LED LED2 and associated with operation of thesecond switching element B2. A low level of the curve cPWM1 or cPWM2corresponds to a closed switch B1 or a closed switch. B2 respectively,i.e. to the current flowing through the corresponding switching elementand bypassing the corresponding LED, which is then turned off. A highlevel of the curve cPWM1 or cPWM2 corresponds to an open switch B1 or anopen switch B2 respectively, i.e. to the current flowing through thecorresponding LED, which is then turned on.

The curves start at a first load condition LC1 during a first phase pLC1in which switch B1 and switch B2 are both closed and the green LED LED1and the blue LED LED2 are both off.

During a second phase pLC2 with a second load condition LC2, switch B1and switch B2 are both open and the green LED LED1 and the blue LED LED2are both on.

During a third phase pLC3 with a third load condition LC3, switch B1 isopen and switch B2 is closed, such that the green LED LED1 is on and theblue LED LED2 is off.

A fourth phase pLC4 is again at load condition LC1, corresponding to thesetting of the switches B1, B2 as in the first phase pLC1.

Another possible load condition LC4 (not shown) corresponds to acondition in which switch B1 is closed and switch B2 is open, such thatthe green LED LED1 is off and the blue LED LED2 is on.

A third curve cVS1 shows the converter current sensing voltage VS as afunction of time when a circuit arrangement according to the prior artis used to regulate the flow of the current through the LED circuitarrangement. An oscillation of the current sensing voltage VS isobserved with a saw-tooth behaviour between an upper trip voltage VH0and a lower trip voltage VL0 during the first phase pLC1, the secondphase pLC2, the third phase pLC3 and the fourth phase pLC4. It may beobserved that the period of the oscillation of the current sensingvoltage VS, and hence also of the current level of the converter currentIL, corresponds to a relatively long duration TLC1 during the firstphase pLC1, to a relatively short duration TLC2 during the second phasepLC2, and to an approximately middle duration TLC3 during the thirdphase pLC3. Load condition LC5 would correspond to another duration TLC5(not shown).

A fourth curve cVSW1 shows the switching control voltage VSW which isoutputted by the hysteretic comparator HCOMP when the circuitarrangement according to the prior art is used to regulate the flow ofthe current through the LED circuit arrangement. The switching controlvoltage VSW is at a logical low level LLL during part of the oscillationwith an a increase time duration in which the converter current sensingvoltage VS is increasing, and is at a logical high level LHL during asecond part of the oscillation with a decrease time duration in whichthe converter current sensing voltage VS is decreasing. The switchingcontrol voltage VSW shows an oscillation with a block-shape behaviourwith the same oscillation periods as the current sensing voltage VS.

A fifth curve cVS2 shows the converter current sensing voltage VS as afunction of time when a circuit arrangement according to the inventionis used to regulate the flow of the current through the LED circuitarrangement. In particular, a circuit arrangement with a switch-modeconverter with a Buck-converter topology similar to the one describedwith reference to FIG. 6 b is used, and a converter current periodcontroller using the switching control voltage VSW for determining theindicator IND associated with the converter current period T is usedsimilar to the one described with reference to FIG. 4 b. Similar curveswould have been obtained when a converter current period controllersimilar to the ones described in reference to FIG. 4 a or 4 c would beused.

An oscillation of the current sensing voltage VS, corresponding to anoscillation of the converter current IL fed to the LED circuitarrangement and hence associated with an oscillation of the currentlevel of the LED current flowing through the LED circuit arrangement, isobserved with a saw-tooth behaviour between an upper trip voltage VH1and a lower trip voltage VL1 during the first phase pLC1. The currentsensing voltage VS oscillates between an upper trip voltage VH2 and alower trip voltage VL2 during the second phase pLC2, with a largeramplitude than during the first phase pLC1, and between an upper tripvoltage VH3 and a lower trip voltage VL3 during the third phase pLC3,with a middle amplitude compared to the amplitude during the first phasepLC1 and the second phase pLC2.

It may be observed that the period of the oscillation of the currentsensing voltage VS, and hence also of the converter current IL,corresponds to a substantially fixed duration, denoted with TCLC1 duringthe first phase pLC1, with TCLC2 during the second phase pLC2, and withTCLC3 during the third phase pLC3.

FIG. 12 shows a further embodiment of the converter current periodcontroller LPCON. The converter current period controller LPCON is inelectrical communication with a memory unit MEM. The memory unit MEMcomprises a table with voltage settings for a plurality of possibleindicator values of the indicator IND, e.g. as illustrated below.

In an example, when applied with the LED circuit arrangement shown inFIG. 10 and discussed in reference with FIG. 11, the plurality ofpossible indicator values are the load conditions LC1, LC2, LC3 and LC4.The load condition may be expressed internally in the circuit and asentries for the table in the memory unit MEM, e.g. as an enumeration ofthe load conditions, as the status of the switches B1, B2 denoted in abinary manner as ‘00’ for LC1, ‘11’ for LC2, ‘10’ for LC3 and ‘01 forLC4’, or in any other suitable manner. The status of the switches B1, B2may be obtained from the LED controller PWMCON.

In an example, the table comprises adjusted upper trip voltage levelsVHA and adjusted lower trip voltage levels VLA, addressed using thebinary notation described above. In one example, the adjusted upper tripvoltage levels VHA and adjusted lower trip voltage levels VLA arepre-determined for each load condition and are fixedly stored in a tablein the memory unit MEM for retrieval only when the load condition ischanged.

In an alternative example, estimates of the adjusted upper trip voltagelevels VHA and adjusted lower trip voltage levels VLA are pre-determinedfor each load condition and are stored in a readable and writable tablein the memory unit MEM for retrieval when the load condition is changed.The converter current period T may then be outside, although in closevicinity, to the period control range Tref. The converter current periodis then measured, e.g. from the current sensing voltage VS as describedabove, and the upper trip voltage level VHA and lower trip voltage levelVLA are adjusted until the converter current period T is inside theperiod control range Tref. The corresponding adjusted upper trip voltagelevel VHA and adjusted lower trip voltage level VLA are then written tothe table in the memory unit MEM. As a result, the system shows aself-learning behaviour in which the voltage level is stored in thememory unit at the latest occurrence of the specific load condition andthe last used voltage level may be retrieved upon a next occurrence ofthe specific load condition.

FIG. 13 shows another further embodiment of the converter current periodcontroller LPCON. The converter current period controller LPCON is inelectrical communication with a spread spectrum generator SSG. Thespread-spectrum generator SSG is operable for varying a centre durationof the period control range Tref over a pre-determined spectral band.This allows to distribute the energy in a well-controlled manner over aspecific spectral band.

FIG. 14 shows an exemplary embodiment of a resistive digital-to-analogueconverter R-DAC usable in embodiments of a circuit arrangement CIRC forestablishing a first voltage V1 and a second voltage V2.

The first voltage V1 and the second voltage V2 may e.g. be used as theupper trip voltage VH and the lower trip voltage VL by the hystereticcomparator HCOMP. Alternatively, the first voltage V1 and the secondvoltage V2 may e.g. be used as the first trip control voltage VC1 andthe second trip control voltage VC2 by the trip control voltagegenerator VCGEN.

As an example, the resistive digital-to-analogue converter R-DAC shownin FIG. 14 comprises a converter reference voltage supply REFSUParranged to provide a converter reference voltage VR. The converterreference voltage supply REFSUP is in electrical communication with aseries circuit of N resistors R1-RN, in the example R1-R6 for N=6.

A first switch array SWA1 comprises a plurality of switches and a secondswitch array SWA2 comprising a plurality of switches. Each plurality ofswitches comprises N+1 switches SWA1.0-SWA1.N, SWA2.0-SWA2.Nrespectively.

Each of the switches SWA1.0-SWA1.N of the first switch array SWA1 is inelectrical communication with the series circuit of resistors R1-RNtapping off at a corresponding position along the series circuit ofresistors R1-RN.

Each switch array SWAT, SWA2 is arranged to be controlled with a digitalcontrol word SWB1, SWB2 comprising a plurality of N+1 bits, the bitsbeing associated with controlling the switches to tap off the seriescircuit of resistors at the corresponding position.

The first voltage V1 and the second voltage V2 are tapped from the sameseries circuit of resistors with the first switch array SWA1 and thesecond switch array SWA2 respectively.

The converter reference voltage supply REFSUP may derive the converterreference voltage VR from, e.g. a bandgap voltage Vbg such that awell-defined voltage reference level is obtained. The bandgap voltageVbg may be amplified by an amplifier A to obtain the converter referencevoltage VR.

When the resistor values for each of the resistors R1-RN are the same,the R-DAC provides a linear voltage divider.

When establishing an adjusted upper trip voltage VHA and an adjustedlower trip voltage VLA from the upper trip voltage VH and the lower tripvoltage VL, the mean voltage level may thus be easily maintained whenstepping up and down the R-DAC with the same number of bits. Theconverter current period T, associated with the difference between theupper trip voltage VH and the lower trip voltage VL, may thus beadjusted while maintaining the mean current level, associated with themean voltage between the upper trip voltage VH and the lower tripvoltage VL.

In an embodiment (not shown), the memory unit MEM described above maycomprise a R-DAC memory RMEM, operable to store and retrieve the digitalcontrol word R-DAC switch settings SWB1, SBW2.

FIG. 15 a schematically shows a circuit composition CC1 a comprising aLED driver IC IC1 a and a LED circuit arrangement LEDCIRC1 a. The LEDdriver IC IC1 a is electrically connected to a LED circuit arrangementLEDCIRC1 a. The LED circuit arrangement LEDCIRC1 a may be a LED circuitarrangement CIRC1 a like the one described in reference with FIG. 10,but may also be another LED arrangement suitable to be driven by the LEDdriver IC IC1 a. The LED driver IC IC1 a comprises an embodiment of acircuit arrangement CIRC according to the invention, comprising aswitch-mode converter SMCONV, a hysteretic comparator HCOMP, a convertercurrent sensor ILSEN and a converter current period controller LPCON.The connections internally in the LED driver IC IC1 a are drawnaccording to an embodiment as shown in FIG. 4 b using the switchingvoltage VSW to determine the indicator associated with the convertercurrent period and employing a Buck-converter topology for the switchmode converter SMCONV like the one discussed in reference with FIG. 6 awith an externally connected inductor L1., but may be according to anysuitable configuration.

The LED driver IC IC1 is connected between a ground voltage GND and aninput voltage Vin. The input voltage Vin is delivered by a power supply(not shown), e.g. a DC power supply delivering a supply voltage of 24 V.

In the example shown, a capacitor Cin1 is placed over the LED driver ICIC1 a to act as a capacitive input filter on the power supply voltageVin.

In the example shown, the LED driver IC IC1 a and the switch-modeconverter SMCONV in the circuit arrangement CIRC1 a are in electricalcommunication with an inductor L1 which is a discrete component externalto the LED driver IC IC1 a. The inductor L1 is in electricalcommunication with the LED circuit arrangement LEDCIRC1 a via aconnection internal in the LED driver IC IC1.

In the example shown, the LED driver IC IC1 a and converter currentsensor ILSEN in the circuit arrangement CIRC1 a are in electricalcommunication with a resistor RS1 which is a discrete component externalto the LED driver IC IC1 a.

A programmable processor uC1, such as a microprocessor, a FPGA, a DSP orany other programmable unit may optionally be connected, as shown by adashed line, to the LED driver IC IC1 a. The processor uC1 mayalternatively or additionally be connected to a LED segment controllerPWMCON1 in the LED circuit arrangement LEDCIRC1 a, as shown by a furtherdashed line.

A computer program product arranged to perform elements of any one ofthe methods implemented as described above, may be loaded in theprogrammable processor, e.g., via an interface connection connectable,directly or via intermediate units, to the programmable processor or toa memory in communication with or included in the programmableprocessor. The computer program product may be read from acomputer-readable medium, e.g., a solid state memory such as a flashmemory, EEPROM, RAM, an optical disk loaded in an optical disk drive, ahard disk drive (HDD), or any other computer-readable medium. Thecomputer-readable medium may be read by a dedicated unit, such as theoptical disk drive to read the optical disk, directly by theprogrammable processor, such as a EEPROM connected to the programmableprocessor, or via other intermediate units.

The programmable processor uC1 may, e.g. comprise a colour controlalgorithm to keep a selected colour balance between the light output ofthe plurality of LEDs.

The programmable processor uC1 may, e.g. cooperate with a LED segmentcontroller PWMCON in the LED circuit arrangement to define the pulsewidth modulation signals.

The programmable processor uC1 may, e.g. cooperate with the spreadspectrum generator SSG to the period control range Tref, or implementthe function of the spectrum generator SSG in the programmable processoruC1 itself.

The programmable processor uC1 may be connected to the LED driver IC IC1as shown in FIG. 15 a. Alternatively, the programmable processor uC1 maybe comprised in the LED driver IC IC1.

The programmable processor uC1 may, e.g. be comprised in the convertercurrent period controller LPCON, to, e.g. implement the determination ofthe indicator and the determination of the trip control voltage valuesin a computer program product. E.g., the programmable processor uC1 maybe arranged to retrieve the status of bypass switches B1, B2 arrangedfor controlling the path of the current ILED1 through a first LED Led1and a second LED Led2 in the LED circuit arrangement LEDCIRC1 a. Thestatus of the switches may be used as a load indicator associated withthe load of the LED circuit arrangement and indicative for the convertercurrent period at a known upper trip voltage VH and lower trip voltageVL. The computer program product loaded in the programmable processoruC1 may be arranged to receive the status of the switches and derive anadjusted upper trip voltage VHA and adjusted lower trip voltage VLA, inorder to obtain a converter current period within the pre-determinedreference window Tref.

The Figure also indicates a further circuit arrangement CIRCINCL whichcan be classified as a circuit arrangement according to the invention.The further circuit arrangement CIRCINCL comprises the LED driver IC IC1a, the optional programmable processor uC1, the inductor L1, theresistor RS1 and the optional capacitor Cin1.

The LED driver IC IC1 a thus provides an integrated circuit whichincludes the circuit to regulate the mean current level and the periodof the converter current ILI.

FIG. 15 b schematically shows a circuit composition CC1 a comprising aLED driver IC IC1 b and a LED circuit arrangement LEDCIRC1 b. The LEDdriver IC IC1 b is electrically connected to a LED circuit arrangementLEDCIRC lb. The LED circuit arrangement LEDCIRC1 b as shown in FIG. 15 bcomprises a series arrangement of a first LED Led1 and a second LEDLed2. The LED driver IC IC1 b comprises an embodiment of a circuitarrangement like described above, comprising a switch-mode converterSMCONV, a hysteretic comparator HCOMP, a converter current sensor ILSEN,a converter current period controller LPCON, and also comprises a LEDsegment controller PWMCON1 b and two bypass switches B1, B2. The LEDsegment controller PWMCON1 b is operable to control two bypass switchesB1, B2. The bypass switch B1 is connected parallel to the first LEDLed1. The bypass switch B2 is connected parallel to the second LED Led2.

The connections internally in the LED driver IC IC1 b are drawn for anexemplary embodiment which uses the status of the bypass switches B1, B2the LED segment as a load indicator communicated from the controllerPWMCON1 b to the converter current period controller LPCON fordetermining the indicator for the converter current period, likediscussed above in reference with FIG. 4 c. The exemplary embodimentemploys a Buck-converter topology for the switch mode converter SMCONVlike the one discussed in reference with FIG. 6 a with an externallyconnected inductor L1. The connections and embodiments of the unitsinside the IC may however alternatively be according to any othersuitable configuration.

The LED driver IC IC1 is connected between a ground voltage GND and aninput voltage Vin. The input voltage Vin is delivered by a power supply(not shown), e.g. a DC power supply delivering a supply voltage of 24 V.

In the example shown, the LED driver IC IC1 a and the switch-modeconverter SMCONV in the circuit arrangement CIRC1 a are in electricalcommunication with an inductor L1 which is a discrete component externalto the LED driver IC IC1 a. The inductor L1 is in electricalcommunication with the LED circuit arrangement LEDCIRC1 a via aconnection internal in the LED driver IC IC1.

In the example shown, the LED driver IC IC1 b and the converter currentsensor ILSEN in the circuit arrangement CIRC1 a are in electricalcommunication with a resistor RS1 which is a discrete component externalto the LED driver IC IC1 b.

A programmable processor uC1, such as a microprocessor, a FPGA, a DSP orany other programmable unit may optionally be connected, as shown by adashed line, to the LED driver IC IC1 b. The processor uC1 communicatewith the LED segment controller PWMCON lb in the LED driver IC IC1 b, asshown by a further dashed line.

The LED driver IC IC1 b thus provides an integrated circuit whichincludes both the circuit to regulate the mean current level and theperiod of the LED current ILED1, and the circuit for operating the LEDswith pulse width modulation. Such an integrated circuit may beappreciated for high-volume applications, as it may provide acost-effective system.

FIG. 16 schematically shows a circuit composition CC2 comprising a LEDdriver IC IC2, a first LED circuit arrangement LEDCIRC1 and a second LEDcircuit arrangement LEDCIRC2. The LED driver IC2 is electricallyconnected to the first LED circuit arrangement LEDCIRC1 and to thesecond LED circuit arrangement LEDCIRC2.

The first LED circuit arrangement LEDCIRC1 may, e.g. be a LED circuitarrangement comprising a green LED Led1 and a blue LED Led2 in series.The second LED circuit arrangement LEDCIRC2 may, e.g. be a LED circuitarrangement comprising a LED segment Led3 comprising two red LEDs and anamber LED Led4 in series.

The LED driver IC2 comprises a first circuit arrangement CIRC 1according to the invention and a second circuit arrangement CIRC2according to the invention, to respectively regulate a first LED currentILED 1 flowing through the first LED circuit arrangement LEDCIRC1 andregulate a second LED current ILED2 flowing through the second LEDcircuit arrangement LEDCIRC2.

The first circuit arrangement CIRC1 and the first LED circuitarrangement LEDCIRC1 are, during use, in electrical communication with afirst inductor L1 and a first resistor Rs1, the first inductor L1 andthe first resistor Rs1 being external to the IC. The second circuitarrangement CIRC2 and the second LED circuit arrangement LEDCIRC2 are,during use, in electrical communication with a second inductor L1 and asecond resistor Rs2, the second inductor L1 and the second resistor Rs2also being external to the IC.

The LED driver IC IC2 further comprises a first LED segment controllerPWMCON1 operable to control two bypass switches B1, B2, also integratedin the IC. The two bypass switches B1, B2 are operable to select thepath of the first LED current ILED1 through the first LED circuitarrangement LEDCIRC 1 and are associated with the green LED Led1 and theblue LED Led2. The bypass switch B1 is connected parallel to the greenLED Led1. The bypass switch B2 is connected parallel to the blue LEDLed2.

The LED driver IC IC2 further comprises a second LED segment controllerPWMCON2 operable to control a further two bypass switches B3, B4, alsointegrated in the IC. The two bypass switches B3, B4 are operable toselect the path of the second LED current ILED2 through the second LEDcircuit arrangement LEDCIRC2 and are associated with the two red LEDsLed3, and the amber LED Led4. The bypass switch B3 is connected parallelto the LED segment Led3, i.e. to the series arrangement of the two redLEDs. The bypass switch B4 is connected parallel to the amber LED Led4.

The first LED segment controller PWMCON1 and the second LED segmentcontroller PWMCON2 may operate each using an individual clock as areference for the pulse width modulation resolution, but mayalternatively be operated from a common clock. When using individualclocks, the clock period associated with the clock may be substantiallythe same or substantially different. In an embodiment, the clockgenerator of the second LED segment controller PWMCON2 behaves as aslave to the first LED segment controller PWMCON1, and the clock of thesecond LED segment controller PWMCON2 is derived from the clock of thefirst LED segment controller PWMCON1. The clocks may be generated in theLED driver IC itself, or be provided from externally, e.g. by anexternally mounted crystal oscillator. The pulse width period may besubstantially the same for the first LED segment controller PWMCON1 andthe second LED segment controller PWMCON2, but may alternatively bedifferent in order not to spread the energy associated with the pulsewidth modulation period over a spectral band.

The LED driver IC IC2 is connected between a ground voltage GND and aninput voltage Vin. The input voltage Vin is delivered by a power supply(not shown), e.g. a DC power supply delivering a supply voltage of 24 V.

The LED driver IC IC2 may be further connected to a programmableprocessor uC2. The programmable processor uC2 may be of similar natureand perform similar functions as the programmable processor uC1described in reference with FIG. 15 a.

The LED driver IC IC2 thus provides an integrated circuit which includesboth the circuit to regulate the mean LED current level and the periodof the oscillation of the LED current, associated with the oscillationof the converter current, and the circuit for operating the LEDs withpulse width modulation, for a lighting system comprising four LEDcolours. The effective light output of each of the four LED colours canbe controlled individually. Hence, a cost-effective lighting system witha high degree of colour control and intensity control may be constructedby employing such an integrated circuit.

FIG. 17 shows an example of a light source 5000 with a LED assembly 4000in a housing 5001. The housing 5001 is a box with, preferably,reflective inner walls. The LED assembly 4000 comprises one or more LEDsand a circuit arrangement employing, during use, one of the methodsimplemented as described above. The light generated by the LED assembly4000 is reflected towards the front of the housing 5001, which iscovered with a diffusive transparent plate 5002. The light source 5000carries a power adapter 5010, which supplies the LED assembly 4000 froma power converter, connected to the mains via a power cord 5011 with apower connecter 5012, to fit a wall contact (not shown) with mainssupply.

It should be noted that the above-mentioned embodiments illustraterather than limit the invention, and that those skilled in the art willbe able to design many alternative embodiments are conceivable withoutdeparting from the scope of the appended claims. E.g., the LED circuitarrangement may comprise more than two segments, each being controllablewith a respective switch, or the LED circuit arrangement may comprise afurther LED segment which is not controllable with a switch, withoutdeparting from the scope of the invention and the appended claims. Inthe claims, any reference signs placed between parentheses shall not beconstrued as limiting the claim.

The invention claimed is:
 1. A method for regulating a LED currentflowing through a LED circuit arrangement at a mean LED current level,the method comprising: establishing a converter current; establishing afirst current control indicator representative of a current level of theconverter current; establishing an oscillation of the converter currentbetween a valley current level and a peak current level in dependence onat least the first current control indicator, wherein the mean LEDcurrent level corresponds to a weighted average of the peak currentlevel and the valley current level of the converter current;establishing a second current control indicator representative of a flowof the converter current; controlling a converter current period of theoscillation of the converter current to be within a period controlrange, the controlling being performed in dependence on at least thesecond current control indicator, and feeding the LED circuitarrangement with at least part of the converter current.
 2. The methodaccording to claim 1, wherein: establishing the first current controlindicator comprises: monitoring a current level of the converter currentand using the monitored current level as the first current controlindicator; establishing the oscillation of the converter currentcomprises: establishing an upper trip current level and a lower tripcurrent level as control crossover thresholds, the upper trip currentlevel being associated with the peak current level of the convertercurrent and the lower trip current level being associated with thevalley current level of the converter current; controlling an increaseof the converter current from a valley current level to a peak currentlevel in response to each crossover of the lower trip current level bythe current level of the converter current in a negative direction, thecontrolling of the increase of the converter current being associatedwith an increase time duration, and controlling a decrease of theconverter current from the peak current level to the valley currentlevel in response to each crossover of the upper trip current level bythe current level of the converter current in a positive direction, thecontrolling of the decrease of the converter current being associatedwith a decrease time duration, and in controlling the converter currentperiod of the oscillation of the converter current, the convertercurrent period corresponds to a sum of the increase time duration andthe decrease time duration.
 3. The method according to claim 1, wherein:establishing the first current control indicator comprises: establishinga converter current sensing voltage representative of the current levelof the converter current and using the converter current sensing voltageas the first current control indicator; establishing the oscillation ofthe converter current comprises: establishing an upper trip voltage anda lower trip voltage as control crossover thresholds, the upper tripvoltage being associated with the peak current level of the convertercurrent and the lower trip voltage being associated with the valleycurrent level of the converter current; controlling an increase of theconverter current from the valley current level to the peak currentlevel in response to each crossover of the lower trip voltage by thesupply current sensing voltage in a negative direction, the controllingof the increase of the converter current being associated with anincrease time duration, and controlling a decrease of the convertercurrent from the peak current level to the valley current level inresponse to each crossover of the upper trip voltage by the convertercurrent sensing voltage in a positive direction, the controlling of thedecrease of the converter current being associated with a decrease timeduration, and in controlling the converter current period of theoscillation of the converter current, the converter current periodcorresponds to a sum of the increase time duration and the decrease timeduration.
 4. The method according to claim 3, further comprising:determining an adjusted upper trip voltage value and an adjusted lowertrip voltage value in dependence on the converter current period,establishing the upper trip voltage and the lower trip voltage from theadjusted upper trip voltage value and the adjusted lower trip voltagevalue, and wherein controlling the converter current period to be withinthe period control range is associated with using the adjusted uppertrip voltage value and the adjusted lower trip voltage in controllingthe increase of the converter current and controlling the decrease ofthe converter current.
 5. The method according to claim 4, whereindetermining the adjusted upper trip voltage and the adjusted lower tripvoltage comprises retrieving at least one voltage-related value from amemory.
 6. The method according to claim 4, wherein the adjusted uppertrip voltage and the adjusted lower trip voltage are determined from theupper trip voltage, the lower trip voltage and an adjustment voltage. 7.The method according to claim 1, wherein establishing the second currentcontrol indicator comprises measuring a current period duration of theconverter current period, and wherein the measured current periodduration is used as the second current control indicator in controllingthe converter current period.
 8. The method according to claim 1,wherein establishing the second current control indicator comprisesdetermining a load associated with the LED circuit arrangement, andwherein the load is used the second current control indicator incontrolling the converter current period.
 9. The method according toclaim 8, wherein establishing the second current control indicatorcomprises determining an estimate of a converter current period durationof the converter current period from the load, and wherein the estimateof the converter current period duration is used as the second currentcontrol indicator in controlling the converter current period.
 10. Themethod according to claim 1, wherein the period control range is definedwith a lower period threshold and an upper period threshold, and thelower duration threshold and the upper duration threshold are determinedfrom a centre duration and a duration width, wherein the duration widthis smaller than 10% of the centre duration.
 11. The method according toclaim 10, wherein the centre duration has a constant value.
 12. Themethod according to claim 10, wherein the centre duration is varied overa spectral band.
 13. The method according to claim 1, furthercomprising: controlling a path of the LED current flowing through theLED circuit arrangement, wherein the LED circuit arrangement comprises afirst LED segment and at least a second LED segment, the first LEDsegment being associated with a first switching element operable forcontrolling a path of the LED current through the first LED segment, thesecond LED segment being associated with a second switching elementoperable for controlling a path of the LED current through the secondLED segment.
 14. The method according to claim 13, wherein the firstswitching element is electrically parallel to the first LED segment, thesecond switching element is electrically parallel to the second LEDsegment, and the first and second switching elements are each operatedto select the path of the LED current to pass through the LED segmentassociated with the respective switching element or to bypass the LEDsegment associated with the respective switching element.
 15. The methodaccording to claim 13, wherein establishing the second current controlindicator comprises determining a load associated with the LED circuitarrangement, wherein the load is used the second current controlindicator in controlling the converter current period, and wherein theload is derived from a status of the first and second switchingelements.
 16. The method according to claim 15, further comprising:storing a hysteresis voltage for the status of the first and secondswitching elements in a memory unit before a change of status of atleast one of the first and second switching elements, and retrieving thehysteresis voltage for the status of the first and second switchingelements from the memory unit after a change of status of at least oneof the first and second switching elements.
 17. The method according toclaim 2, wherein: establishing the first current control indicatorcomprises: establishing a converter current sensing voltagerepresentative of the current level of the converter current and usingthe converter current sensing voltage as the first current controlindicator; establishing the oscillation of the converter currentcomprises: establishing an upper trip voltage and a lower trip voltageas control crossover thresholds, the upper trip voltage being associatedwith the peak current level of the converter current and the lower tripvoltage being associated with the valley current level of the convertercurrent; controlling an increase of the converter current from thevalley current level to the peak current level in response to eachcrossover of the lower trip voltage by the supply current sensingvoltage in a negative direction, the controlling of the increase of theconverter current being associated with an increase time duration, andcontrolling a decrease of the converter current from the peak currentlevel to the valley current level in response to each crossover of theupper trip voltage by the converter current sensing voltage in apositive direction, the controlling of the decrease of the convertercurrent being associated with a decrease time duration, and incontrolling the converter current period of the oscillation of theconverter current, the converter current period corresponds to a sum ofthe increase time duration and the decrease time duration.
 18. Themethod according to claim 17, further comprising: determining anadjusted upper trip voltage value and an adjusted lower trip voltagevalue in dependence on the converter current period, establishing theupper trip voltage and the lower trip voltage from the adjusted uppertrip voltage value and the adjusted lower trip voltage value, andwherein controlling the converter current period to be within the periodcontrol range is associated with using the adjusted upper trip voltagevalue and the adjusted lower trip voltage in controlling the increase ofthe converter current and controlling the decrease of the convertercurrent.
 19. The method according to claim 18, further comprising:determining an adjusted upper trip voltage value and an adjusted lowertrip voltage value in dependence on the converter current period,establishing the upper trip voltage and the lower trip voltage from theadjusted upper trip voltage value and the adjusted lower trip voltagevalue, and wherein controlling the converter current period to be withinthe period control range is associated with using the adjusted uppertrip voltage value and the adjusted lower trip voltage in controllingthe increase of the converter current and controlling the decrease ofthe converter current.
 20. The method according to claim 19, whereindetermining the adjusted upper trip voltage and the adjusted lower tripvoltage comprises retrieving at least one voltage-related value from amemory.